Hybrid digital-analog mmwave repeater/relay with full duplex

ABSTRACT

A system for transmitting millimeter wave signals includes a plurality of transceivers for communicating with a plurality of remote locations over millimeter wave communications links. Each of the plurality of transceivers further includes a patch antenna array having a plurality of patch antennas. The plurality of patch antennas includes a transmitter array portion in a first orientation for transmitting signals and a receiver array portion in a second orientation for receiving signals. The first and second orientations limit interference between the transmitted signals and the received signals. Baseband processing circuitry converts between millimeter wave and baseband signals. The plurality of transceivers relays the millimeter wave signals between at least a first millimeter wave transceiver at first one of the plurality of remote locations and a second millimeter wave transceiver at a second one of the plurality of remote locations.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of U.S. Patent Application No.62/965,019, filed Jan. 23, 2020, entitled HYBRID DIGITAL-ANALOG MMWAVEREPEATER/RELAY WITH FULL DUPLEX, which is incorporated by referenceherein in its entirety.

TECHNICAL FIELD

The present invention related to repeater/relay transmitters, and moreparticularly, to a millimeter wave repeater/relay using full duplex.

SUMMARY

The present invention, as disclosed and described herein, in one aspectthereof, comprises a system for transmitting millimeter wave signalsincludes a plurality of transceivers for communicating with a pluralityof remote locations over millimeter wave communications links. Each ofthe plurality of transceivers further includes a patch antenna arrayhaving a plurality of patch antennas. The plurality of patch antennasincludes a transmitter array portion in a first orientation fortransmitting signals and a receiver array portion in a secondorientation for receiving signals. The first and second orientationslimit interference between the transmitted signals and the receivedsignals. Baseband processing circuitry converts between millimeter waveand baseband signals. The plurality of transceivers relays themillimeter wave signals between at least a first millimeter wavetransceiver at first one of the plurality of remote locations and asecond millimeter wave transceiver at a second one of the plurality ofremote locations.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding, reference is now made to thefollowing description taken in conjunction with the accompanyingDrawings in which:

FIG. 1 is a block diagram of a building penetration system;

FIG. 2 illustrates the bi-directional nature of the building penetrationsystem for transmissions from the outside;

FIG. 3 illustrates the bi-directional nature of the building penetrationsystem for transmissions from the inside;

FIG. 4 illustrates a network deployment of the building penetrationsystem of FIG. 1A;

FIG. 5 illustrates a block diagram of an optical bridge for transmittingmillimeter wave transmissions through a window;

FIG. 6 is a more detailed block diagram of the millimeter waveregeneration and retransmission circuitry;

FIG. 7 illustrates the RF transceiver circuitry of the millimeter waveregeneration and retransmission circuitry;

FIG. 8 illustrates various techniques for increasing spectral efficiencywithin a transmitted signal;

FIG. 9 illustrates a particular technique for increasing spectralefficiency within a transmitted signal;

FIG. 10 illustrates a general overview of the manner for providingcommunication bandwidth between various communication protocolinterfaces;

FIG. 11 illustrates the manner for utilizing multiple level overlaymodulation with twisted pair/cable interfaces;

FIG. 12 illustrates a general block diagram for processing a pluralityof data streams within an optical communication system;

FIG. 13 is a functional block diagram of a system for generating orbitalangular momentum within a communication system;

FIG. 14 is a functional block diagram of the orbital angular momentumsignal processing block of FIG. 7;

FIG. 15 is a functional block diagram illustrating the manner forremoving orbital angular momentum from a received signal including aplurality of data streams;

FIG. 16 illustrates a single wavelength having two quanti-spinpolarizations providing an infinite number of signals having variousorbital angular momentums associated therewith;

FIG. 17A illustrates a plane wave having only variations in the spinangular momentum;

FIG. 17B illustrates a signal having both spin and orbital angularmomentum applied thereto;

FIGS. 18A-18C illustrate various signals having different orbitalangular momentum applied thereto;

FIG. 18D illustrates a propagation of Poynting vectors for various Eigenmodes;

FIG. 18E illustrates a spiral phase plate;

FIG. 19 illustrates various manners for converting a Gaussian beam intoan OAM beam;

FIG. 20A illustrates a fabricated metasurface phase plate;

FIG. 20B illustrates a magnified structure of the metasurface phaseplate;

FIG. 21A illustrates the steps to produce 24 multiplex OAM beams;

FIG. 21B illustrates the optical spectrum of a WDM signal carrier on anOAM beam;

FIG. 22A illustrates a turbulence emulator;

FIG. 22B illustrates the measured power distribution of an OAM beamafter passing through turbulence with a different strength;

FIG. 23A illustrates how turbulence effects mitigation using adaptiveoptics;

FIG. 23B illustrates experimental results of distortion mitigation usingadaptive optics;

FIG. 24 is a functional block diagram of a residential IP networksystem;

FIG. 25 illustrates the manner in which a mmwave system may be utilizedto transmit information to a residential IP network system;

FIG. 26 illustrates a first embodiment for wireless transmission ofbroadband data to a residential IP network system;

FIG. 27 illustrates a second embodiment for wireless transmission ofbroadband data to a residential IP network system;

FIG. 28 illustrates a third embodiment for wireless transmission ofbroadband data to a residential IP network system;

FIG. 29 illustrates an implementation of vOLTHA on an OLT and ONU link;

FIG. 30 illustrates a broadband link between an OLT and home gateway;

FIG. 31 illustrates the interface between and ONU and the plurality ofhome gateways;

FIG. 32 illustrates a first embodiment of a broadband datacommunications link between an OLT and home gateway;

FIG. 33 illustrates a second embodiment of a broadband datacommunications link between an OLT and virtual reality goggles;

FIG. 34 illustrates the structure for communication through a window orwall using transceiver dongles;

FIG. 35A illustrates a functional block diagram of a RK 3399 processorand a Peraso chipset;

FIG. 35B illustrates a more detailed block diagram of a RK 3399processor and a Peraso chipset;

FIG. 35C illustrates a block diagram of a transceiver dongle forproviding full-duplex communications;

FIG. 35D illustrates a block diagram a device implementing a Perasochipset;

FIG. 36 is a top-level block diagram of a Peraso transceiver;

FIGS. 37A and 37B illustrate a detailed application diagram of a Perasochipset;

FIG. 38 illustrates serial transmissions between Peraso transceivers;

FIG. 39 illustrates parallel transmissions between Peraso transceivers;

FIG. 40 illustrates a transceiver dongle and a multilevel patch antennaarray for transmitting and receiving OAM signals;

FIG. 41 illustrates a side view of the transceiver dongle;

FIG. 42 illustrates a transceiver dongle and a multilevel patch antennaarray for transmitting and receiving OAM signals;

FIG. 43 illustrates a view of a three layer patch antenna;

FIG. 44 illustrates a top view of the three layer patch antenna;

FIG. 45 illustrates the separation between antenna layer of the threelayer patch antenna;

FIG. 46 illustrates the analog and digital cancellation for multipletransmit and receive chains;

FIG. 47 illustrates a block diagram of the transmitter and receivercircuit;

FIG. 48 illustrates transmit and receive paths for the analog anddigital cancellation processes.

FIG. 49 illustrates a simplified block diagram of an RF receiver;

FIG. 50 illustrates a block diagram of a full-duplex design with threecancellation techniques;

FIG. 51 illustrates a block diagram of a full-duplex system;

FIG. 52 illustrates a circuit for measuring the cancellation performanceof signal inversion versus offset;

FIG. 53 illustrates the cancellation of a self-interference signal withbalun versus with phase offset;

FIG. 54 illustrates cancellation performance with increasing signalbandwidth using a balun method versus using a phase offset cancellation;

FIG. 55 illustrates full-duplex transmissions between first and secondtransceivers;

FIG. 56 illustrates full-duplex transmissions using orbital angularmomentum between first and second transceivers;

FIG. 57 illustrates signals received by antennas in full-duplex with OAMcommunications;

FIG. 58 illustrates a block diagram of a transceiver implementingfull-duplex communications;

FIG. 59 illustrates a block diagram of a Backhaul Network KeyPerformance Indicator;

FIG. 60 illustrates a small cell private network;

FIG. 61 illustrates a small cell node having a primary link and one ormore backup links;

FIG. 62 illustrates a small cell node including means for multiplexingbetween multiple transceiver types;

FIG. 63 illustrates an SDN-based architecture for link generation;

FIG. 64 illustrates a small cell node implementing fast localrestoration within its data plane layer;

FIG. 65 illustrates a flow diagram describing the process forimplementing SDN-based local repair;

FIG. 66 illustrates a flow diagram describing the process for detectinglink state and for the transmission on primary and backup links;

FIG. 67 illustrates a private network wirelessly connecting to users toprovide a connection to the network core;

FIG. 68 illustrates a mesh network for interconnecting an edge networkwith users;

FIG. 69 illustrates the components of a self-organized network foraccess;

FIG. 70 illustrates a number of mesh network nodes;

FIG. 71 illustrates a mesh network with interfering structure;

FIG. 72 illustrates a mesh network of access sites connected to multipleusers;

FIG. 73 illustrates the wireless connections between an optimallylocated access site and an associated network site and resident areenterprise users;

FIG. 74 illustrates a network of a plurality of optimally located accesssites and network sites using both licensed and unlicensed data bands;

FIG. 75 illustrates the software components of a 60 GHz last dropsystem;

FIG. 76 illustrates the manner in which a central controller and meshclient operate on top of a WiGig baseband layer;

FIG. 77 illustrates the various mesh software functions for the meshcontroller and mesh node;

FIG. 78 illustrates a system providing full cancellation operation;

FIG. 79 illustrates a full-duplex cancellation technique;

FIG. 80 illustrates vertically separated patch antennas;

FIG. 81 illustrates combination of self interference cancellationtechniques with OAM orthogonality techniques;

FIG. 82 illustrates a first manner for improving isolation usingorthogonal Gaussian modes;

FIG. 83 illustrates a second matter for improving isolation usingorthogonal Gaussian modes;

FIG. 84 illustrates a table of effects caused by rotation of a circularreceiver array within a patch antenna;

FIG. 85 illustrates a patch antenna array having a 12 patch antennacircular receiver antenna array;

FIG. 86 illustrates a table of effects caused by rotation of a circularreceiver of Ray including 12 patch antennas;

FIG. 87 illustrates vertically separated transmitter arrays and receiverarrays;

FIG. 88 illustrates received voltage values for different patch antennaconfigurations;

FIGS. 89-106 illustrate various voltage plots for differing patchantenna array durations of a circular transmitter array and a circularreceiver array;

FIG. 107 illustrates a for transmitting and receiving system;

FIG. 108 illustrates various transmission towers for antenna;

FIG. 109 illustrates a link budget plot for a patch antenna array;

FIG. 110 a transmitting and receiving patch antenna arrays;

FIG. 111 illustrates a link budget for transmitting signals in crossissues for a transmitting and receiving array;

FIG. 112 illustrates a block diagram of a system for transmittingmillimeter wave signals;

FIG. 113 illustrates various functionalities associated with arepeater/relay;

FIG. 114 illustrates a single relay system;

FIG. 115 illustrates a tandem relay configuration;

FIG. 116 illustrates a wide junction relay configuration;

FIG. 117 illustrates an H junction relay configuration;

FIG. 118 illustrates a tandem wide junction relay configuration;

FIG. 119 illustrates a configuration of tandem repeaters and a passivereflector;

FIG. 120 illustrates the full duplex and half duplex communications usedwithin a Peter system;

FIG. 121 illustrates a block diagram of a transceiver dongleimplementing the millimeter wave repeater/relay; and

FIG. 122 illustrates the various functionalities implemented within themillimeter/wave repeater/relay.

DETAILED DESCRIPTION

Referring now to the drawings, wherein like reference numbers are usedherein to designate like elements throughout, the various views andembodiments of hybrid digital-analog mmwave repeater/relay with fullduplex and various embodiments associated therewith are illustrated anddescribed, and other possible embodiments are described. The figures arenot necessarily drawn to scale, and in some instances the drawings havebeen exaggerated and/or simplified in places for illustrative purposesonly. One of ordinary skill in the art will appreciate the many possibleapplications and variations based on the following examples of possibleembodiments.

One issue with wireless telecommunications is the inability of highfrequency RF waves to penetrate through windows and walls of homes andbusiness offices. If a window includes any infrared (IR) shielding inorder to save energy within the house or office building, the losses insignals transmitted through the shielding are typically up to 40 or 50dB. Thus, the millimeter wave system described herein provides theability to provide tunneling of such optical and high frequency radiowaves without requiring the need to drill through the glass, window orbuilding to provide a physical portal therethrough would provide greatbenefits to wireless communication technologies. This may be done at anyfrequency that has problems penetrating through the glass or building.Glass is one of the most popular and versatile due to its constantlyimproving solar and thermal performance. One manner for achieving thisperformance is through the use of passive and solar control lowemissivity coatings. These low emissivity glass materials produce a hugeloss for millimeter wave spectrum transmissions and create a hugeproblem for transmission of millimeter waves through such glass. Thesystem described herein below provides for the ability to allowfrequencies having a problem penetrating through a glass or building tobe processed in such a manner to enable the signals to be transmittedinto or out of a home or building.

Millimeter wave signaling was developed when the FCC devised a band planmaking 1300 MHz of the local multipoint distribution service (LMDS)spectrum available within each basic trading area across the UnitedStates. The plan allocated two LMDS licenses per BTA (basic tradingarea), an “A Block” and a “B Block” in each. The A Block licensecomprised 1150 MHz of total bandwidth, and the B Block license consistedof 150 MHz of total bandwidth. A license holder Teligent developed asystem for fixed wireless point to multipoint technology that could sendhigh speed broadband from rooftops to surrounding small and medium-sizebusinesses. However, the system, as well as others provided by Winstarand NextLink, did not succeed and many of the LMDS licenses fell backinto the hands of the FCC. These licenses and related spectrum are seenas useful for 5G trials and services.

Additionally there is a need to provide signals between the base stationtransceivers that are connected to a voice or data transmission networkand the devices associated with the homes or buildings enabling thetransmission of signals into the interior of the buildings as describedmore fully herein.

Referring now to FIG. 1, there is illustrated a general block diagram ofthe building penetration transmission system 102. The buildingpenetration transmission system 102 uses 5G fixed millimeter wavedeployments to overcome high building penetration losses due to RF andoptical obstructions such as windows, brick and concrete walls. Thebuilding penetration transmission system 102 greatly increases thenumber of enterprise and residential buildings where 5G millimeter wavesignals can be used to deliver gigabyte ethernet services. The systemprovides an optical or RF tunnel through the window or wall 106 withoutrequiring the drilling of any holes or the creation of some type ofsignal permeable portal within the window or wall. The generation ofdirection radio waves using the describe system enables the generationof directional beams to tunnel through low-e glass or walls. The systemenables link budgets between the interior and exterior transceivers besatisfied. The system greatly increases the number of building that mayuse millimeter wave signals to deliver Gigabit Ethernet using consumerinstalled devices.

The building penetration transmission system 102 generally includes anexterior repeater transmitter 104 located on the exterior of the windowor wall 106. The repeater transmitter 104 transmits and receives anumber of frequencies including 2.5 GHz band, 3.5 GHz band, 5 GHz band,24 GHz band, 28 GHz band (A1, A2, B1 and B2), 39 GHz band, 60 GHz band,71 GHz band and 81 GHz band. The 3.5 GHz band is CBRS (Citizens BandRadio Service), the 60 GHz band is V-band and the 71 GHz and 81 GHz areE-band. The repeater transmitter 104 is powered using magnetic resonanceor inductive coupling such that the outside unit requires no externalpower source. The repeater 104 transmits received signals through thewindow or wall 106 to a transceiver 108 located on the interior of thebuilding. The transceiver 108 includes an antenna 110 for providingethernet and/or power connections. The building penetration transmissionsystem 102 may provide one gigabit per second throughput traffictunneling through a building structure such as a window or wall. Thetransceiver 108 may include a port 112 providing femto cellconnectivity, but in general transmits Wi-Fi indoors using the antenna110. Alternatively, the ethernet or power connections can be hardwiredto the transceiver 108. The building penetration transmission system 102may be located at any point on a wall or window of a structure. Thebuilding penetration transmission system 102 is designed to work withdifferent types of walls and windows in order to enable millimeter wavesignals to penetrate different types of structures. The repeater 104 andtransceiver 108 are constructed of a metal/plastic design to withstandthe harshest environments including precipitation, hot/cold weather andhigh/low humidity.

The transceiver 108 includes gigabyte ethernet ports, a power output, atleast one USB 2.0 port and dual flash image support. The buildingpenetration transmission system 102 provides a range of up to 200 feet(60 m). The system requires a 24 V/M passive gigabyte PoE and has a 20 Wmaximum power consumption that may be powered using magnetic resonancewireless charging in one embodiment. The system provides 2 GHz ofchannel bandwidth 60 GHz.

FIGS. 2 and 3 illustrates the bidirectional communication betweentransceiver 104 located on the exterior side of the window or wall 106and transceiver 108 located on the interior side of the window or wall106. A remote base station transmitter 109 transmits wireless signals toan external transceiver 104. Communication transmissions from theexterior transceiver 104 to the interior transceiver 108 occur over acommunications link 114. The signals transmitted to the interior maythen be transmitted to consumer premises equipment (CPE) 111 using beamforming or WiFi 113 from an internal router 115. As shown in FIG. 1C,internal devices 117 (such as mobile devices or Internet-of-Thingsdevices) transmit signals to the internal router 115. The internalrouter 115 provides the signals to the internal transceiver 108.Transmissions from the interior of the window or wall 106 to theexterior are from transceiver 108 to transceiver 104 are oncommunications link 116. The external transceiver 104 then transmits thesignals to the external base station 109. Thus, the system enablesbidirectional communications that may utilize RF, optical or other typesof communication technologies as more fully described hereinbelow.

Referring now to FIG. 4, there is illustrated a network deployment ofthe building penetration system discussed with respect to FIGS. 1-3. Aprovider network 130 interfaces with the local network through fiber PoP(point of presence) cabinets 132. The cabinets 132 have a fiber link 134to an access point 136. Each of the access points 136 wirelesslycommunicates with a network of other access points 136 that are locatedfor example on light poles within a local area over wirelesscommunication links using any number of communication frequencies aswill be described herein. The access points 136 communicate withtransceiver systems 138 that comprise the building penetration systemdescribed herein where in signals are wirelessly transmitted to anexternal transceiver and then transmitted to the interior of thebusiness or home such that information may be bi-directionallytransmitted from the provider network 130 to/from devices located withinthe interior of various structures. In this manner, data may be providedbetween the network provider 130 and devices of all types located withinthe structures using wireless communications that normally would notpenetrate to the interior of the structures due to losses occurring bypenetration of the signals into the interior of the structures.

FIG. 5 illustrates one manner for transmitting millimeter wave signalsinside of a building using an optical bridge 502 mounted to a window504. The optical bridge 502 includes a first portion 506 included on anoutside of the window 504 and a second portion 508 included on theinside of the window 504. The first portion 506 includes a 28 GHztransceiver 510 that is mounted on the outside of the window 504. The 28GHz transceiver 510 receives the millimeter wave transmissions that arebeing transmitted from, for example, a base station 104 such as thatdescribed with respect to FIG. 1. The received/transmitted signals aretransmitted to and from the transceiver 510 using a receiver opticalsubassembly (ROSA)/transmission optical subassembly (TOSA) 512. Areceiver optical subassembly is a component used for receiving opticalsignals in a fiber optic system. Similarly, a transceiver opticalsubassembly is a component used for transmitting optical signals in afiber optic system. ROSA/TOSA component 512 transmits or receives theoptical signals through the window 504 to a ROSA/TOSA component 514located on the inside of the window 504. The signals are forwarded fromthe ROSA/TOSA 514 to a Wi-Fi transmitter 516 for transmissions withinthe building.

Referring now to FIG. 6, there is illustrated a more detailedillustration of the components for transmitting millimeter wavetransmissions through a window or wall of a building. The transceiver600 includes an optional antenna gain element 602 for receiving themillimeter wave transmissions transmitted on a down/up link 604 from abase station 605. The down/uplink 604 comprises a 28 GHz beamtransmission. However other frequency transmissions may also beutilized. An RF receiver 606 is used for receiving information from thebase station 204 over the down/up link 604. Similarly, the RFtransmitter 608 is used for transmitting information on the down/up link604 to a base station 605. Receive signals are provided to a demodulator610 for demodulation of any received signals. The demodulated signalsare provided to a groomer 612 which places the signals in theappropriate configuration for transmission by the optical transmissioncomponents. When translating different modulations (say from a highorder QAM to OOK (On-Off Keying)), there are signaling conversions thatrequire some grooming (or signal conditioning) to ensure all bitstranslate properly and still provide a low BER. The present systemtranslates from RF at a high QAM rate to raw bit rates of OOK to enabletransmissions using the VCSELs to go through the glass of the window.VCSELs only work with OOK and therefore a translation using the groomer612 is needed. If a received signal were just down-convert from 28 GHzdirectly to 5.8 GHz (because 5.8 GHz does pass through the wall andglass), then we do not need to worry about complications of translatingto low order modulation. The problem is that down-converting signal from28 GHz to 5.8 GHz requires expensive components. The groomer 612completes the translation of the received 28 GHz signal to a frequencyfor transmission through a glass or wall without the more expensivecomponents.

The signals to be transmitted are passed through an amplifier 614 toamplify the signal for transmission. The amplified signal is provided toVCSELs 616 for optically transmitting the signal. The VCSEL 616 is avertical cavity surface emitting laser that is a type of semiconductorlaser diode with laser beam omissions perpendicular from the topsurface. In a preferred embodiment, the VCSEL 616 comprises a FinisarVCSEL having a wavelength of approximately 780 nm, a modulation rate of4 Gb per second and an optical output power of 2.2 mW (3.4 to dBm). Inalternative embodiments the components for transmitting the opticalsignals across the window 604 may comprise an LED (light emitting diode)or EEL (edge emitting lasers). The different lasers enable differentoptical re-transmissions at different frequencies based on differentcharacteristics of a window such as tint.

The VCSEL 616 includes a transmission optical subassembly (TOSA) forgenerating the optical signals for transmission from VCSEL 616 to VCSEL618 located on the opposite side of the window 604. The VCSELs 616 and618 comprise a laser source for generating the optical signals fortransmission across the window 604. In one embodiment, the VCSELcomprises a Finisar VCSEL that provides a 780 nm optical signal having amaximum modulation rate of 4 Gb per second when running at 1 Gb persecond and an optical output power of 3 mW (5 dBm). The TOSA includes alaser device or LED device for converting electrical signals from theamplifier 614 into light signal transmissions. Transmissions from theoutside VCSEL 616 to the inside VCSEL 418 and an associated receiveroptical subassembly (ROSA).

The optical signals are transmitted through the window 604 using opticalfocusing circuitry 617. The optical link 628 between VCSEL 616 and VCSEL618 has an optical link budget associated therewith that defines thelosses that may be accepted while still transmitting the informationbetween the VCSELs 616, 618. The VCSEL has an output power ofapproximately 5 dBm. The detector at the receiver within the VCSEL candetect a signal at approximately −12 dBm. The glass losses associatedwith the optical signal passing through the glass at a wavelength of 780nm is 7.21 dB. The coupling loss and lens gain associated with thetransmission is approximately 0.1 dB. The maximum displacement losscaused by a lens displacement of 3.5 mm is 6.8 dB. Thus, the total linkmargin equals 2.88 dB based upon a subtraction of the detectorsensitivity, glass losses, coupling loss and lens gain and maximumdisplacement loss from the VCSEL output power. The 2.88 dB link marginis provided for unexpected an extra losses such as len's losses andunexpected output variances.

Referring now to FIG. 7 there is illustrated a more detailed blockdiagram of the transceiver 310. The receiver portion 702 includes an RFreceiver 704 for receiving the RF signals transmitted from the basestation on the downlink 706. The receiver 704 generates output signalshaving a real portion BBI 708 and an imaginary portion BBQ 710. The RFreceiver 704 generates the real signal 708 and imaginary signal 710responsive to the receive signal and inputs from a phase lockedloop/voltage control oscillator 705. The phase locked loop/voltagecontrol oscillator 705 provides inputs to the RF receiver 704 responsiveto a reference oscillator signal provided from reference oscillator 707and a voltage controlled oscillator signal provided from oscillator 709.The real signal 708 and the imaginary signal 710 are provided toanalog-to-digital converters 712 for conversion to a digital signal. Theanalog-to-digital converters 712 are clocked by an associated clockinput 714 provided from clock generation circuit 716. The clockgeneration circuit 716 also receives an input from the referenceoscillator 707. The real and imaginary digital signals 718 and 720 areinput to a digital down converter 722. The digital signals are downconverted to a lower frequency and output as a bit stream 724 to theoptical transmission circuitry (VCSEL) for transmitting across thewindow glass.

The transmitter portion 724 receives a digital bitstream 726 from theoptical circuitry and provides this bitstream to the real and imaginaryportions of digital up converters 728 to convert the digital data to ahigher frequency for transmission. The real and imaginary portions ofthe up-converted digital signal are provided to a crest factor reductionprocessor 730. Some signals (especially OFDM-based systems) have highpeak-to-average power ratio (PAR) that negatively impacts the efficiencyof power amplifiers (PAs). Crest factor reduction (CFR) schemesimplemented by the processor help reduce PAR and have been used for manynetworks (CDMA & OFDM). However, CFR schemes developed primarily forCDMA signals have a poor performance when used in in OFDM (given thetight error vector magnitude (EVM) requirements). With a well-designedCFR algorithm on FPGAs, one can achieve low-latency, high-performancethat can significantly reduce the PAR of the output signal whichimproves PA efficiency and reduced cost.

The real and imaginary signals are provided from the crest factorreduction processor 730 to a digital to analog converter 732. Thedigital to analog converter 732 converts the real and imaginary digitalsignals into real and imaginary analog signals BBI 734 and BBQ 736. Thereal and imaginary analog signals are inputs to the RF transmitter 738.The RF transmitter 738 processes the real signal 734 and imaginarysignal 736 responsive to input from the phase locked loop/voltagecontrol oscillator 704 to generate RF signals for transmission on theuplink 740 to generate the millimeter wave and transmissions.

Referring now to FIG. 8, the communication system configurationintroduces two techniques, one from the signal processing techniques 804category and one from the creation of new eigen channels 806 categorythat are entirely independent from each other. Their combinationprovides a unique manner to disrupt the access part of an end to endcommunications system from twisted pair and cable to fiber optics, tofree space optics, to RF used in cellular, backhaul and satellite. Thefirst technique involves the use of a new signal processing techniqueusing new orthogonal signals to upgrade QAM modulation using nonsinusoidal functions. This particular embodiment is referred to asquantum level overlay (QLO) 902 as shown in FIG. 9. The secondembodiment involves the application of new electromagnetic wavefrontsusing a property of electromagnetic waves or photon, called orbitalangular momentum (QAM) 904. Application of each of the quantum leveloverlay techniques 902 and orbital angular momentum application 904uniquely offers orders of magnitude higher spectral efficiency 906within communication systems in their combination.

With respect to the quantum level overlay technique 902, new eigenfunctions are introduced that when overlapped (on top of one anotherwithin a symbol) significantly increases the spectral efficiency of thesystem. The quantum level overlay technique 302 borrows from quantummechanics, special orthogonal signals that reduce the time bandwidthproduct and thereby increase the spectral efficiency of the channel.Each orthogonal signal is overlaid within the symbol acts as anindependent channel. These independent channels differentiate thetechnique from existing modulation techniques.

With respect to the application of orbital angular momentum 904, thisembodiment introduces twisted electromagnetic waves, or light beams,having helical wave fronts that carry orbital angular momentum (OAM).Different OAM carrying waves/beams can be mutually orthogonal to eachother within the spatial domain, allowing the waves/beams to beefficiently multiplexed and demultiplexed within a communications link.OAM beams are interesting in communications due to their potentialability in special multiplexing multiple independent data carryingchannels.

With respect to the combination of quantum level overlay techniques 902and orbital angular momentum application 904, the combination is uniqueas the OAM multiplexing technique is compatible with otherelectromagnetic techniques such as wave length and polarization divisionmultiplexing. This suggests the possibility of further increasing systemperformance. The application of these techniques together in highcapacity data transmission disrupts the access part of an end to endcommunications system from twisted pair and cable to fiber optics, tofree space optics, to RF used in cellular/backhaul and satellites.

Each of these techniques can be applied independent of one another, butthe combination provides a unique opportunity to not only increasespectral efficiency, but to increase spectral efficiency withoutsacrificing distance or signal to noise ratios.

Using the Shannon Capacity Equation, a determination may be made ifspectral efficiency is increased. This can be mathematically translatedto more bandwidth. Since bandwidth has a value, one can easily convertspectral efficiency gains to financial gains for the business impact ofusing higher spectral efficiency. Also, when sophisticated forward errorcorrection (FEC) techniques are used, the net impact is higher qualitybut with the sacrifice of some bandwidth. However, if one can achievehigher spectral efficiency (or more virtual bandwidth), one cansacrifice some of the gained bandwidth for FEC and therefore higherspectral efficiency can also translate to higher quality.

Telecom operators and vendors are interested in increasing spectralefficiency. However, the issue with respect to this increase is thecost. Each technique at different layers of the protocol has a differentprice tag associated therewith. Techniques that are implemented at aphysical layer have the most impact as other techniques can besuperimposed on top of the lower layer techniques and thus increase thespectral efficiency further. The price tag for some of the techniquescan be drastic when one considers other associated costs. For example,the multiple input multiple output (MIMO) technique uses additionalantennas to create additional paths where each RF path can be treated asan independent channel and thus increase the aggregate spectralefficiency. In the MIMO scenario, the operator has other associated softcosts dealing with structural issues such as antenna installations, etc.These techniques not only have tremendous cost, but they have hugetiming issues as the structural activities take time and the achievingof higher spectral efficiency comes with significant delays which canalso be translated to financial losses.

The quantum level overlay technique 902 has an advantage that theindependent channels are created within the symbols without needing newantennas. This will have a tremendous cost and time benefit compared toother techniques. Also, the quantum layer overlay technique 902 is aphysical layer technique, which means there are other techniques athigher layers of the protocol that can all ride on top of the QLOtechniques 902 and thus increase the spectral efficiency even further.QLO technique 902 uses standard QAM modulation used in OFDM basedmultiple access technologies such as WIMAX or LTE. QLO technique 902basically enhances the QAM modulation at the transceiver by injectingnew signals to the I & Q components of the baseband and overlaying thembefore QAM modulation as will be more fully described herein below. Atthe receiver, the reverse procedure is used to separate the overlaidsignal and the net effect is a pulse shaping that allows betterlocalization of the spectrum compared to standard QAM or even the rootraised cosine. The impact of this technique is a significantly higherspectral efficiency.

Referring now more particularly to FIG. 10, there is illustrated ageneral overview of the manner for providing improved communicationbandwidth within various communication protocol interfaces 1002, using acombination of multiple level overlay modulation 1004 and theapplication of orbital angular momentum 1006 to increase the number ofcommunications channels. The following discussions of orbital angularmomentum processing and multiple level overlay modulation illustrate twotechniques that may or may not be implemented in RF transmissions in thebelow described systems and embodiments. RF transmissions may beconfigured to implement one, both or neither of the techniques in thedescribed embodiments.

The various communication protocol interfaces 1002 may comprise avariety of communication links, such as RF communication, wirelinecommunication such as cable or twisted pair connections, or opticalcommunications making use of light wavelengths such as fiber-opticcommunications or free-space optics. Various types of RF communicationsmay include a combination of RF microwave or RF satellite communication,as well as multiplexing between RF and free-space optics in real time.

By combining a multiple layer overlay modulation technique 1004 withorbital angular momentum (OAM) technique 1006, a higher throughput overvarious types of communication links 1002 may be achieved. The use ofmultiple level overlay modulation alone without OAM increases thespectral efficiency of communication links 1002, whether wired, optical,or wireless. However, with OAM, the increase in spectral efficiency iseven more significant.

Multiple overlay modulation techniques 1004 provide a new degree offreedom beyond the conventional 2 degrees of freedom, with time T andfrequency F being independent variables in a two-dimensional notationalspace defining orthogonal axes in an information diagram. This comprisesa more general approach rather than modeling signals as fixed in eitherthe frequency or time domain. Previous modeling methods using fixed timeor fixed frequency are considered to be more limiting cases of thegeneral approach of using multiple level overlay modulation 1004. Withinthe multiple level overlay modulation technique 1004, signals may bedifferentiated in two-dimensional space rather than along a single axis.Thus, the information-carrying capacity of a communications channel maybe determined by a number of signals which occupy different time andfrequency coordinates and may be differentiated in a notationaltwo-dimensional space.

Within the notational two-dimensional space, minimization of the timebandwidth product, i.e., the area occupied by a signal in that space,enables denser packing, and thus, the use of more signals, with higherresulting information-carrying capacity, within an allocated channel.Given the frequency channel delta (Δf), a given signal transmittedthrough it in minimum time Δt will have an envelope described by certaintime-bandwidth minimizing signals. The time-bandwidth products for thesesignals take the form;ΔtΔf=½(2n+1)where n is an integer ranging from 0 to infinity, denoting the order ofthe signal.

These signals form an orthogonal set of infinite elements, where eachhas a finite amount of energy. They are finite in both the time domainand the frequency domain, and can be detected from a mix of othersignals and noise through correlation, for example, by match filtering.Unlike other wavelets, these orthogonal signals have similar time andfrequency forms.

The orbital angular momentum process 1006 provides a twist to wavefronts of the electromagnetic fields carrying the data stream that mayenable the transmission of multiple data streams on the same frequency,wavelength, or other signal-supporting mechanism. This will increase thebandwidth over a communications link by allowing a single frequency orwavelength to support multiple eigen channels, each of the individualchannels having a different orthogonal and independent orbital angularmomentum associated therewith.

Referring now to FIG. 11, there is illustrated a further communicationimplementation technique using the above described techniques as twistedpairs or cables carry electrons (not photons). Rather than using each ofthe multiple level overlay modulation 1004 and orbital angular momentumtechniques 1006, only the multiple level overlay modulation 1004 can beused in conjunction with a single wireline interface and, moreparticularly, a twisted pair communication link or a cable communicationlink 1102. The operation of the multiple level overlay modulation 1104,is similar to that discussed previously with respect to FIG. 10, but isused by itself without the use of orbital angular momentum techniques1006, and is used with either a twisted pair communication link or cableinterface communication link 1102.

Referring now to FIG. 12, there is illustrated a general block diagramfor processing a plurality of data streams 1202 for transmission in anoptical communication system. The multiple data streams 1202 areprovided to the multi-layer overlay modulation circuitry 1204 whereinthe signals are modulated using the multi-layer overlay modulationtechnique. The modulated signals are provided to orbital angularmomentum processing circuitry 1206 which applies a twist to each of thewave fronts being transmitted on the wavelengths of the opticalcommunication channel. The twisted waves are transmitted through theoptical interface 1208 over an optical communications link such as anoptical fiber or free space optics communication system. FIG. 12 mayalso illustrate an RF mechanism wherein the interface 1208 wouldcomprise and RF interface rather than an optical interface.

Referring now more particularly to FIG. 13, there is illustrated afunctional block diagram of a system for generating the orbital angularmomentum “twist” within a communication system, such as that illustratedwith respect to FIG. 10, to provide a data stream that may be combinedwith multiple other data streams for transmission upon a same wavelengthor frequency. Multiple data streams 1302 are provided to thetransmission processing circuitry 1300. Each of the data streams 1302comprises, for example, an end to end link connection carrying a voicecall or a packet connection transmitting non-circuit switch packed dataover a data connection. The multiple data streams 1302 are processed bymodulator/demodulator circuitry 1304. The modulator/demodulatorcircuitry 1304 modulates the received data stream 1302 onto a wavelengthor frequency channel using a multiple level overlay modulationtechnique, as will be more fully described herein below. Thecommunications link may comprise an optical fiber link, free-spaceoptics link, RF microwave link, RF satellite link, wired link (withoutthe twist), etc.

The modulated data stream is provided to the orbital angular momentum(OAM) signal processing block 1306. Each of the modulated data streamsfrom the modulator/demodulator 1304 are provided a different orbitalangular momentum by the orbital angular momentum electromagnetic block1306 such that each of the modulated data streams have a unique anddifferent orbital angular momentum associated therewith. Each of themodulated signals having an associated orbital angular momentum areprovided to an optical transmitter 1308 that transmits each of themodulated data streams having a unique orbital angular momentum on asame wavelength. Each wavelength has a selected number of bandwidthslots B and may have its data transmission capability increase by afactor of the number of degrees of orbital angular momentum 1 that areprovided from the OAM electromagnetic block 1306. The opticaltransmitter 1308 transmitting signals at a single wavelength couldtransmit B groups of information. The optical transmitter 1308 and OAMelectromagnetic block 1306 may transmit 1×B groups of informationaccording to the configuration described herein.

In a receiving mode, the optical transmitter 1308 will have a wavelengthincluding multiple signals transmitted therein having different orbitalangular momentum signals embedded therein. The optical transmitter 1308forwards these signals to the OAM signal processing block 1306, whichseparates each of the signals having different orbital angular momentumand provides the separated signals to the demodulator circuitry 1304.The demodulation process extracts the data streams 1302 from themodulated signals and provides it at the receiving end using themultiple layer overlay demodulation technique.

Referring now to FIG. 14, there is provided a more detailed functionaldescription of the OAM signal processing block 1406. Each of the inputdata streams are provided to OAM circuitry 1402. Each of the OAMcircuitry 1402 provides a different orbital angular momentum to thereceived data stream. The different orbital angular momentums areachieved by applying different currents for the generation of thesignals that are being transmitted to create a particular orbitalangular momentum associated therewith. The orbital angular momentumprovided by each of the OAM circuitries 1402 are unique to the datastream that is provided thereto. An infinite number of orbital angularmomentums may be applied to different input data streams using manydifferent currents. Each of the separately generated data streams areprovided to a signal combiner 1404, which combines the signals onto awavelength for transmission from the transmitter 1406.

Referring now to FIG. 15, there is illustrated the manner in which theOAM processing circuitry 1306 may separate a received signal intomultiple data streams. The receiver 1502 receives the combined OAMsignals on a single wavelength and provides this information to a signalseparator 1504. The signal separator 1504 separates each of the signalshaving different orbital angular momentums from the received wavelengthand provides the separated signals to OAM de-twisting circuitry 1506.The OAM de-twisting circuitry 1506 removes the associated OAM twist fromeach of the associated signals and provides the received modulated datastream for further processing. The signal separator 1504 separates eachof the received signals that have had the orbital angular momentumremoved therefrom into individual received signals. The individuallyreceived signals are provided to the receiver 1502 for demodulationusing, for example, multiple level overlay demodulation as will be morefully described herein below.

FIG. 16 illustrates in a manner in which a single wavelength orfrequency, having two quanti-spin polarizations may provide an infinitenumber of twists having various orbital angular momentums associatedtherewith. The 1 axis represents the various quantized orbital angularmomentum states which may be applied to a particular signal at aselected frequency or wavelength. The symbol omega (ω) represents thevarious frequencies to which the signals of differing orbital angularmomentum may be applied. The top grid 1602 represents the potentiallyavailable signals for a left handed signal polarization, while thebottom grid 1604 is for potentially available signals having righthanded polarization.

By applying different orbital angular momentum states to a signal at aparticular frequency or wavelength, a potentially infinite number ofstates may be provided at the frequency or wavelength. Thus, the stateat the frequency Δω or wavelength 1606 in both the left handedpolarization plane 1602 and the right handed polarization plane 1604 canprovide an infinite number of signals at different orbital angularmomentum states Δl. Blocks 1608 and 1610 represent a particular signalhaving an orbital angular momentum Δl at a frequency Δω or wavelength inboth the right handed polarization plane 1604 and left handedpolarization plane 1610, respectively. By changing to a differentorbital angular momentum within the same frequency Δω or wavelength1606, different signals may also be transmitted. Each angular momentumstate corresponds to a different determined current level fortransmission from the optical transmitter. By estimating the equivalentcurrent for generating a particular orbital angular momentum within theoptical domain and applying this current for transmission of thesignals, the transmission of the signal may be achieved at a desiredorbital angular momentum state.

Thus, the illustration of FIG. 16, illustrates two possible angularmomentums, the spin angular momentum, and the orbital angular momentum.The spin version is manifested within the polarizations of macroscopicelectromagnetism, and has only left and right hand polarizations due toup and down spin directions. However, the orbital angular momentumindicates an infinite number of states that are quantized. The paths aremore than two and can theoretically be infinite through the quantizedorbital angular momentum levels.

Using the orbital angular momentum state of the transmitted energysignals, physical information can be embedded within the radiationtransmitted by the signals. The Maxwell-Heaviside equations can berepresented as:

${\nabla{\cdot E}} = \frac{\rho}{ɛ_{0}}$${\nabla{\times E}} = {- \frac{\partial B}{\partial t}}$ ∇⋅B = 0${\nabla{\times B}} = {{ɛ_{0}\mu_{0}\frac{\partial E}{\partial t}} + {\mu_{0}{j\left( {t,x} \right)}}}$where ∇ is the del operator, E is the electric field intensity and B isthe magnetic flux density. Using these equations, one can derive 23symmetries/conserved quantities from Maxwell's original equations.However, there are only ten well-known conserved quantities and only afew of these are commercially used. Historically if Maxwell's equationswhere kept in their original quaternion forms, it would have been easierto see the symmetries/conserved quantities, but when they were modifiedto their present vectorial form by Heaviside, it became more difficultto see such inherent symmetries in Maxwell's equations.

Maxwell's linear theory is of U(1) symmetry with Abelian commutationrelations. They can be extended to higher symmetry group SU(2) form withnon-Abelian commutation relations that address global (non-local inspace) properties. The Wu-Yang and Harmuth interpretation of Maxwell'stheory implicates the existence of magnetic monopoles and magneticcharges. As far as the classical fields are concerned, these theoreticalconstructs are pseudo-particle, or instanton. The interpretation ofMaxwell's work actually departs in a significant ways from Maxwell'soriginal intention. In Maxwell's original formulation, Faraday'selectrotonic states (the Aμ field) was central making them compatiblewith Yang-Mills theory (prior to Heaviside). The mathematical dynamicentities called solitons can be either classical or quantum, linear ornon-linear and describe EM waves. However, solitons are of SU(2)symmetry forms. In order for conventional interpreted classicalMaxwell's theory of U(1) symmetry to describe such entities, the theorymust be extended to SU(2) forms.

Besides the half dozen physical phenomena (that cannot be explained withconventional Maxwell's theory), the recently formulated Harmuth Ansatzalso address the incompleteness of Maxwell's theory. Harmuth amendedMaxwell's equations can be used to calculate EM signal velocitiesprovided that a magnetic current density and magnetic charge are addedwhich is consistent to Yang-Mills filed equations. Therefore, with thecorrect geometry and topology, the Aμ potentials always have physicalmeaning

The conserved quantities and the electromagnetic field can berepresented according to the conservation of system energy and theconservation of system linear momentum. Time symmetry, i.e. theconservation of system energy can be represented using Poynting'stheorem according to the equations:

$H = {{\sum\limits_{i}{m_{i}\gamma_{i}c^{2}}} + {\frac{ɛ_{0}}{2}{\int{d^{3}{x\left( {{E}^{2} + {c^{2}{B}^{2}}} \right)}}}}}$

-   -   Hamiltonian (total energy)

${\frac{dU^{mech}}{dt} + \frac{dU^{em}}{dt} + {\oint_{s^{\prime}}{d^{2}x^{\prime}{\hat{n^{\prime}} \cdot S}}}} = 0$

-   -   conservation of energy

The space symmetry, i.e., the conservation of system linear momentumrepresenting the electromagnetic Doppler shift can be represented by theequations:

$p = {{\sum\limits_{i}{m_{i}\gamma_{i}v_{i}}} + {ɛ_{0}{\int{d^{3}{x\left( {E \times B} \right)}}}}}$

-   -   linear momentum

${\frac{{dp}^{mech}}{dt} + \frac{{dp}^{em}}{dt} + {\oint_{s^{\prime}}{d^{2}x^{\prime}{\hat{n^{\prime}} \cdot T}}}} = 0$

-   -   conservation of linear momentum        The conservation of system center of energy is represented by        the equation:

$R = {{\frac{1}{H}{\sum\limits_{i}{\left( {x_{i} - x_{0}} \right)m_{i}\gamma_{i}c^{2}}}} + {\frac{ɛ_{0}}{2H}{\int{d^{3}{x\left( {x - x_{0}} \right)}\left( {{E^{2}} + {c^{2}{B^{2}}}} \right)}}}}$

Similarly, the conservation of system angular momentum, which gives riseto the azimuthal Doppler shift is represented by the equation:

${\frac{dJ^{mech}}{dt} + \frac{dJ^{em}}{dt} + {\oint_{s^{\prime}}{d^{2}x^{\prime}{\hat{n^{\prime}} \cdot M}}}} = 0$

-   -   conservation of angular momentum        For radiation beams in free space, the EM field angular momentum        J^(em) can be separated into two parts:        J ^(em)=ε₀∫_(V′) d ³ x′(E×A)+ε₀∫_(V′) d ³ E _(i)[(x′−x ₀)×∇]A        _(i)        For each singular Fourier mode in real valued representation:

$J^{em} = {{{- i}\frac{ɛ_{0}}{2\omega}{\int_{V^{\prime}}{d^{3}{x^{\prime}\left( {E^{*} \times E} \right)}}}} - {i\frac{ɛ_{0}}{2\omega}{\int_{V^{\prime}}{d^{3}x^{\prime}{E_{i}\left\lbrack {\left( {x^{\prime} - x_{0}} \right) \times \nabla} \right\rbrack}E_{i}}}}}$

The first part is the EM spin angular momentum Sem, its classicalmanifestation is wave polarization. And the second part is the EMorbital angular momentum Lem its classical manifestation is wavehelicity. In general, both EM linear momentum Pem, and EM angularmomentum Jem=Lem+Sem are radiated all the way to the far field.

By using Poynting theorem, the optical vorticity of the signals may bedetermined according to the optical velocity equation:

${{\frac{\partial U}{\partial t} + {\nabla{\cdot S}}} = 0},$

-   -   continuity equation        where S is the Poynting vector

${S = {\frac{1}{4}\left( {{E \times H^{*}} + {E^{*} \times H}} \right)}},$and U is the energy density

${u = {\frac{1}{4}\left( {{ɛ{E}^{2}} + {\mu_{0}{H}^{2}}} \right)}},$with E and H comprising the electric field and the magnetic field,respectively, and ε and μ₀ being the permittivity and the permeabilityof the medium, respectively. The optical vorticity V may then bedetermined by the curl of the optical velocity according to theequation:

$V = {{\nabla \times v_{opt}} = {\nabla \times \left( \frac{{E \times H^{*}} + {E^{*} \times H}}{{ɛ{E}^{2}} + {\mu_{0}{H}^{2}}} \right)}}$

Referring now to FIGS. 17A and 17B, there is illustrated the manner inwhich a signal and its associated Poynting vector in a plane wavesituation. In the plane wave situation illustrated generally at 1702,the transmitted signal may take one of three configurations. When theelectric field vectors are in the same direction, a linear signal isprovided, as illustrated generally at 1704. Within a circularpolarization 1706, the electric field vectors rotate with the samemagnitude. Within the elliptical polarization 1708, the electric fieldvectors rotate but have differing magnitudes. The Poynting vectorremains in a constant direction for the signal configuration to FIG. 17Aand always perpendicular to the electric and magnetic fields. Referringnow to FIG. 17B, when a unique orbital angular momentum is applied to asignal as described here and above, the Poynting vector S 1710 willspiral about the direction of propagation of the signal. This spiral maybe varied in order to enable signals to be transmitted on the samefrequency as described herein.

FIGS. 18A through 18C illustrate the differences in signals havingdifferent helicity (i.e., orbital angular momentums). Each of thespiraling Poynting vectors associated with the signals 1802, 1804, and1806 provide a different shaped signal. Signal 1802 has an orbitalangular momentum of +1, signal 1804 has an orbital angular momentum of+3, and signal 1806 has an orbital angular momentum of −4. Each signalhas a distinct angular momentum and associated Poynting vector enablingthe signal to be distinguished from other signals within a samefrequency. This allows differing type of information to be transmittedon the same frequency, since these signals are separately detectable anddo not interfere with each other (Eigen channels).

FIG. 18D illustrates the propagation of Poynting vectors for variousEigen modes. Each of the rings 1820 represents a different Eigen mode ortwist representing a different orbital angular momentum within the samefrequency. Each of these rings 1820 represents a different orthogonalchannel. Each of the Eigen modes has a Poynting vector 1822 associatedtherewith.

Topological charge may be multiplexed to the frequency for either linearor circular polarization. In case of linear polarizations, topologicalcharge would be multiplexed on vertical and horizontal polarization. Incase of circular polarization, topological charge would multiplex onleft hand and right hand circular polarizations. The topological chargeis another name for the helicity index “I” or the amount of twist or OAMapplied to the signal. The helicity index may be positive or negative.In RF, different topological charges can be created and muxed togetherand de-muxed to separate the topological charges.

The topological charges 1 s can be created using Spiral Phase Plates(SPPs) as shown in FIG. 18E using a proper material with specific indexof refraction and ability to machine shop or phase mask, hologramscreated of new materials or a new technique to create an RF version ofSpatial Light Modulator (SLM) that does the twist of the RF waves (asopposed to optical beams) by adjusting voltages on the device resultingin twisting of the RF waves with a specific topological charge. SpiralPhase plates can transform a RF plane wave (1=0) to a twisted RF wave ofa specific helicity (i.e. 1=+1).

Cross talk and multipath interference can be corrected using RFMultiple-Input-Multiple-Output (MIMO). Most of the channel impairmentscan be detected using a control or pilot channel and be corrected usingalgorithmic techniques (closed loop control system).

As described previously with respect to FIG. 13, each of the multipledata streams applied within the processing circuitry has a multiplelayer overlay modulation scheme applied thereto.

Mode Conversion Approaches

Referring now to FIG. 19, among all external-cavity methods, perhaps themost straightforward one is to pass a Gaussian beam through a coaxiallyplaced spiral phase plate (SPP) 1902. An SPP 1902 is an optical elementwith a helical surface, as shown in FIG. 18E. To produce an OAM beamwith a state of f, the thickness profile of the plate should be machinedas lλθ/2π(n−1), where n is the refractive index of the medium. Alimitation of using an SPP 1902 is that each OAM state requires adifferent specific plate. As an alternative, reconfigurable diffractiveoptical elements, e.g., a pixelated spatial light modulator (SLM) 1904,or a digital micro-mirror device can be programmed to function as anyrefractive element of choice at a given wavelength. As mentioned above,a helical phase profile exp(ilθ) converts a linearly polarized Gaussianlaser beam into an OAM mode, whose wave front resembles an l-foldcorkscrew 1906, as shown at 1904. Importantly, the generated OAM beamcan be easily changed by simply updating the hologram loaded on the SLM1904. To spatially separate the phase-modulated beam from thezeroth-order non-phase-modulated reflection from the SLM, a linear phaseramp is added to helical phase code (i.e., a “fork”-like phase pattern1908 to produce a spatially distinct first-order diffracted OAM beam,carrying the desired charge. It should also be noted that theaforementioned methods produce OAM beams with only an azimuthal indexcontrol. To generate a pure LG_(l,p) mode, one must jointly control boththe phase and the intensity of the wavefront. This could be achievedusing a phase-only SLM with a more complex phase hologram.

Some novel material structures, such as metal-surface, can also be usedfor OAM generation. A compact metal-surface could be made into a phaseplate by manipulation of the structure caused spatial phase response. Asshown in FIGS. 20A and 20B, a V-shaped antenna array 2002 is fabricatedon the metal surface 2004, each of which is composed of two arms 2006,2008 connected at one end 2010. A light reflected by this plate wouldexperience a phase change ranging from 0 to 2π, determined by the lengthof the arms and angle between two arms. To generate an OAM beam, thesurface is divided into 8 sectors 2012, each of which introduces a phaseshift from 0 to 7π/4 with a step of π/4. The OAM beam with l=+1 isobtained after the reflection, as shown in FIG. 20C.

A following experiment doubled the spectral efficiency by adding thepolarization multiplexing into the OAM-multiplexed free-space data link.Four different OAM beams (l=+4, +8, −8, +16) on each of two orthogonalpolarizations (eight channels in total) were used to achieve a Terabit/stransmission link. The eight OAM beams were multiplexed anddemultiplexed using the same approach as mentioned above. The measuredcrosstalk among channels carried by the eight OAM beams is shown inTable 1, with the largest crosstalk being ˜−18.5 dB. Each of the beamswas encoded with a 42.8 Gbaud 16-QAM signal, allowing a total capacityof ˜1.4 (42.8×4×4×2) Tbit/s.

TABLE 1 OAM₊₄ OAM₊₈ OAM⁻⁸ OAM₊₁₆ Measured Crosstalk X- Y- X- Y- X- Y- X-Y- OAM₊₄ (dB) X-Pol −23 2 −26.7 −30.8 −30.5 −27.7 −24.6 −30.1 Y-Pol−25.7 OAM₊₈ (dB) X-Pol −26.6 −23.5 −21.6 −18.9 −25.4 −23.9 −28.8 Y-Pol−25.0 OAM⁻⁸ (dB) X-Pol −27.5 −33.9 −27.6 −30.8 −20.5 −26.5 −21.6 Y-Pol−26.8 OAM₊₁₆ (dB) X-Pol −24.5 −31.2 −23.7 −23.3 −25.8 −26.1 −30.2 Y-Pol−24.0 Total from other OAMs * −21.8 −21.0 −21.2 −21.4 −18.5 −21.2 −22.2−20.7

The capacity of the free-space data link was further increased to 100Tbit/s by combining OAM multiplexing with PDM (phase divisionmultiplexing) and WDM (wave division multiplexing). In this experiment,24 OAM beams (l=±4, ±7, ±10, ±13, ±16, and ±19, each with twopolarizations) were prepared using 2 SLMs, the procedures for which areshown in FIG. 21A at 2102-2106. Specifically, one SLM generated asuperposition of OAM beams with l=+4, +10, and +16, while the other SLMgenerated another set of three OAM beams with l=+7, +13, and +19 (FIG.21A). These two outputs were multiplexed together using a beam splitter,thereby multiplexing six OAM beams: l=+4, +7, +10, +13, +16, and +19(FIG. 21A). Secondly, the six multiplexed OAM beams were split into twocopies. One copy was reflected five times by three mirrors and two beamsplitters, to create another six OAM beams with inverse charges (FIG.21B). There was a differential delay between the two light paths tode-correlate the data. These two copies were then combined again toachieve 12 multiplexed OAM beams with l=±4, ±7, ±10, ±13, ±16, and ±19(FIG. 21B). These 12 OAM beams were split again via a beam splitter. Oneof these was polarization-rotated by 90 degrees, delayed by ˜33 symbols,and then recombined with the other copy using a polarization beamsplitter (PBS), finally multiplexing 24 OAM beams (with l=±4, ±7, ±10,±13, ±16, and ±19 on two polarizations). Each of the beam carried a WDMsignal comprising 100 GHz-spaced 42 wavelengths (1,536.34-1,568.5 nm),each of which was modulated with 100 Gbit/s QPSK data. The observedoptical spectrum of the WDM signal carried on one of the demultiplexedOAM beams (l=+10).

Atmospheric Turbulence Effects on OAM Beams

One of the critical challenges for a practical free-space opticalcommunication system using OAM multiplexing is atmospheric turbulence.It is known that inhomogeneities in the temperature and pressure of theatmosphere lead to random variations in the refractive index along thetransmission path, and can easily distort the phase front of a lightbeam. This could be particularly important for OAM communications, sincethe separation of multiplexed OAM beams relies on the helicalphase-front. As predicted by simulations in the literature, theserefractive index inhomogeneities may cause inter-modal crosstalk amongdata channels with different OAM states.

The effect of atmospheric turbulence is also experimentally evaluated.For the convenience of estimating the turbulence strength, one approachis to emulate the turbulence in the lab using an SLM or a rotating phaseplate. FIG. 22A illustrates an emulator built using a thin phase screenplate 2202 that is mounted on a rotation stage 2204 and placed in themiddle of the optical path. The pseudo-random phase distributionmachined on the plate 2202 obeys Kolmogorov spectrum statistics, whichare usually characterized by a specific effective Fried coherence lengthr0. The strength of the simulated turbulence 4906 can be varied eitherby changing to a plate 2202 with a different r0, or by adjusting thesize of the beam that is incident on the plate. The resultant turbulenceeffect is mainly evaluated by measuring the power of the distorted beamdistributed to each OAM mode using an OAM mode sorter. It was foundthat, as the turbulence strength increases, the power of the transmittedOAM mode would leak to neighboring modes and tend to be equallydistributed among modes for stronger turbulence. As an example, FIG. 22Bshows the measured average power (normalized) 1=3 beam under differentemulated turbulence conditions. It can be seen that the majority of thepower is still in the transmitted OAM mode 2208 under weak turbulence,but it spreads to neighboring modes as the turbulence strengthincreases.

Turbulence Effects Mitigation Techniques

One approach to mitigate the effects of atmospheric turbulence on OAMbeams is to use an adaptive optical (AO) system. The general idea of anAO system is to measure the phase front of the distorted beam first,based on which an error correction pattern can be produced and can beapplied onto the beam transmitter to undo the distortion. As for OAMbeams with helical phase fronts, it is challenging to directly measurethe phase front using typical wavefront sensors due to the phasesingularity. A modified AO system can overcome this problem by sending aGaussian beam as a probe beam to sense the distortion, as shown in FIG.23A. Due to the fact that turbulence is almost independent of the lightpolarization, the probe beam is orthogonally polarized as compared toall other beams for the sake of convenient separation at beam separator2302. The correction phase pattern can be derived based on the probebeam distortion that is directly measured by a wavefront sensor 2304. Itis noted that this phase correction pattern can be used tosimultaneously compensate multiple coaxially propagating OAM beams. FIG.23B at 2310-2320 illustrate the intensity profiles of OAM beams with1=1, 5 and 9, respectively, for a random turbulence realization with andwithout mitigation. From the far-field images, one can see that thedistorted OAM beams (upper), up to 1=9, were partially corrected, andthe measured power distribution also indicates that the channelcrosstalk can be reduced.

Another approach for combating turbulence effects is to partially movethe complexity of optical setup into the electrical domain, and usedigital signal processing (DSP) to mitigate the channel crosstalk. Atypical DSP method is the multiple-input-multiple-output (MIMO)equalization, which is able to blindly estimate the channel crosstalkand cancel the interference. The implementation of a 4×4 adaptive MIMOequalizer in a four-channel OAM multiplexed free space optical linkusing heterodyne detection may be used. Four OAM beams (1=+2, +4, +6 and+8), each carrying 20 Gbit/s QPSK data, were collinearly multiplexed andpropagated through a weak turbulence emulated by the rotating phaseplate under laboratory condition to introduce distortions. Afterdemultiplexing, four channels were coherently detected and recordedsimultaneously. The standard constant modulus algorithm is employed inaddition to the standard procedures of coherent detection to equalizethe channel interference. Results indicate that MIMO equalization couldbe helpful to mitigate the crosstalk caused by either turbulence orimperfect mode generation/detection, and improve both error vectormagnitude (EVM) and the bit-error-rate (BER) of the signal in anOAM-multiplexed communication link. MIMO DSP may not be universallyuseful as outage could happen in some scenarios involving free spacedata links. For example, the majority power of the transmitted OAM beamsmay be transferred to other OAM states under a strong turbulence withoutbeing detected, in which case MIMO would not help to improve the systemperformance.

FIG. 24 illustrates a functional block diagram of a residential IPnetwork system 2402. An input 2404 from a millimeter wave transmissionsystem that enables the transmission of millimeter waves from anexterior transmission unit to the interior of the structure providesbroadband signals to a residential IP network gateway 2406. Theresidential IP network gateway 2406 determines where the signal comingfrom the input 2404 needs to be routed and provides the output on one ofa plurality of possible outputs to the appropriate destination IPaddress associated with the device requesting the broadband information.The output lines may comprise a coaxial cable 2408, an ethernet cable2410 or existing phone line 2412. The coaxial cables 2408 may provideinputs to a set top box 2414 that then provides an output to a livingroom TV 2416 through for example an HDMI connection 2418. A firstethernet connection 2410 may connect to a set top box/DVR 2420. Afurther ethernet connection 2422 provides data to a second television2424. Ethernet connections 2410 may also provide data to a PC 2426 or anetwork drive 2428. The existing phone line connection 2412 would beprovided to a phone outlet 2430 for connection of a telephone. Finally,a Wi-Fi antenna 2432 provides the ability for the residential IP networkGateway 2462 to provide a Wi-Fi network connection within a structure.The Wi-Fi network connection enables devices such as a PC 2434, laptop2436, iPad 2438 or iPhone 2440 to wirelessly connect to the residentialIP network Gateway 2406 to receive broadband data.

FIG. 25 illustrates the manner in which a millimeter wave system may beutilized to transmit information to a residential IP network system. Anaccess unit 2502 located on the outside of a structure wirelesslytransmits broadband data to CPE (customer premises equipment) units 2504located within one or more structures associated with the residential IPnetwork system. The access unit 2502 may receive the broadband data fortransmission to the CPE units 2504 via wireless transmissions or ahardwired connection. The wireless access provided between the accessunit 2502 and the CPE units 2504 may be provided in any of a number offrequency bands including, but not limited to millimeter bands 24 GHz,28 GHz, 39 GHz, 60 GHz as well as 2.5 GHz, the CBRS band 3.5 GHz, Wi-Fibands at 2.4 and 5 GHz. The signals are transmitted from outside thestructure to inside the structure using any of the above describedtransmission techniques for transmitting signals through a wall orwindow. Within the structure the CPE unit 2504 uses Wi-Fi or otherunlicensed bands within the premises to transmit signals to Internet ofthings (TOT) devices 2506, PCs 2508, IP TVs 2510, closed circuittelevisions 2512, IP telephones 2514 and Wi-Fi extenders 2516. These areonly some examples of IP-based devices and any type of Wi-Fi connectabledevice may be utilized within the structure for communications with theCPE 2504. The manner in which broadband data may be transmitted from theexterior of the structure to the interior of the structure may beconfigured utilizing the above described millimeter wave transmissionsystems in a number of fashions.

FIG. 26 illustrates a first embodiment wherein the access unit 2502wirelessly transmits the broadband data to a millimeter wave systemtransceiver 2602 located on an external side of a window or wall 2604.The system is consumer installed with the repeater (transceiver 2602)outside of the building and a transceiver 2606 on the inside of thebuilding. This configuration uses millimeter wave transmitters on bothsides of the glass or wall enabling tunneling of radio waves usingeither optical signals or RF signals. The broadband signals areconnected directly to the CPE device 2504 via electronic integration atan integrated window unit to provide access to the residential IPnetwork 2608. The wireless transmissions to the millimeter wavetransceiver 2602 may be within any frequency band including, but notlimited to, millimeter wave bands such as to 24 GHz, 28 GHz, 39 GHz, 60GHz and 2.5 GHz; CBRS bands such as 3.5 GHz; and Wi-Fi bands such as 2.4and 5 GHz. Millimeter wave transceiver 2602 transmits the signalsthrough the window or wall 2604 to a second millimeter wave transceiver2606 located on the interior of the structure. The composition of themillimeter wave transceivers 2602 and 2606 may be any of those discussedherein above with respect to systems for transmitting signals through awindow or wall 2604. The interior millimeter wave transceiver 2606outputs received data to (or receives data from) a customer premisesequipment 2504 associated with the residential network IP 2608. Themillimeter wave transceiver 2606 and CPE 2504 may comprise integratedequipment within a same box or device for receiving the signals from themillimeter wave transceiver 2602 located on the external of thestructure and providing the data to the residential IP network 2608 andassociated devices.

FIG. 27 illustrates an alternative embodiment wherein the access unit2502 wirelessly transmits the broadband data signals to the externalmillimeter wave transceiver 2602 as described previously with respect toFIG. 26. In this embodiment, millimeter wave transceivers are providedon sides of the window or wall 2604 enabling tunneling of radio wavesusing either optical signals or RF signals. The signals transmittedthrough the window or wall 2604 are then wirelessly connected to the CPE2504 using either an unlicensed band or unlicensed Wi-Fi withbeamforming. The external millimeter wave transceiver 2602 transmits thedata through the window or wall 2604 as described herein to an internalmillimeter wave transceiver 2606. The internal millimeter wavetransceiver 2606 incorporates a beamforming device or Wi-Fi router thatallows for transmission of the received signals using beam forminglicense or Wi-Fi to an integrated millimeter wave transceiver 2501 andCPE 2504. The CPE 2504 transmits the data to the residential IP network2608 and associated devices.

Referring now to FIG. 28 there is illustrated a further embodiment of asystem for transmitting the broadband signals to a residential IPnetwork 2608 wherein the access unit 2402 wirelessly transmits thesignals to a millimeter wave transceiver 2802 located on an externalside of a window 2804 of a building or structure. A millimeter wavetransceiver 2802 is located on the outside of a window glass and useshigh power phased array and beamforming circuitry 2803 to enabletunneling of radio waves to wirelessly connect to the CPE 2808 located adistance from the window 2804 using either a licensed band or unlicensedWi-Fi. The millimeter wave transceiver 2802 includes a high-poweredphased array 2803 providing beamforming or Wi-Fi router capabilities fortransmitting signals wirelessly through the window 2804 to a millimeterwave transceiver 2806 located on an interior of the structure but placedat a location that is not directly on the opposite side of the window2804. The millimeter wave transceiver 2806 is integrated with the CPE2808 that transmits the broadband data to the residential IP network2608 and associated devices.

The described system provides an optical or RF tunnel that allowssignals to be transmitted from outside a building to devices within thebuilding. Once the broadband access is delivered into the premises(residential or commercial), other unlicensed bands can be used insidethe premises. The optical or RF tunnel can also be used to allow signalsfrom the Internet of Things devices located within the building to gofrom inside to outside. In addition to the techniques described hereinabove, other near field techniques can be used for transmitting theinformation through the window or wall.

Millimeter Wave with Optical Networks

Each access technology brings its own protocols and concepts which meanscontrol and management of legacy access devices can be a problem. vOLTHAconfines the differences of access technology to the locality of accessand hides them from the upper layers of the OSS stack. Referring now toFIG. 29 there is illustrated the implementation of vOLTHA with an OLT2902 and ONU 2904 link. The OLT 2902 communicates with multiple ONUs2904 through a splitter 2905. vOLTHA containers communicate over gRPC.The main container publishes events to Kafka and persists data in Consulfor service discovery. Southbound OLT adapters 2906 and ONU adapters2908 will be their own containers as well. OLT adapter 2906 and ONUadapter 2908 enables OLT-ONU interoperability through the vOLTHA core29310. The ONU adapter 2908 sends OMCI (ONT management controlinterface) to the OLT 2902 through the OLT adapter 2906.

Using vOLTHA to create hardware abstraction layers for Wave Agilityenables integration to a residential network IP gateway over mmWaveFixed Wireless Access (gigabit rate access with Dynamic QoS-Application& Network slicing support). One of the challenges faced in the nextgeneration broadband access at gigabit rates is the need for runningfiber to the home or business. Referring now to FIG. 30, with fixedmmWave 5G wireless access technology ONUs 12402 (PON end points) can beutilized for the aggregation of self-installed fixed wireless accesspoints.

Almost all recent FTTH (fiber to the home) deployments, as well as thosecurrently being planned, use passive optical networking. The concept ofa Passive Optical Network (PON) 3012, involves the use of passive fibersplitters which allow multiple customers (typically 32-128) to share asingle fiber pair. GPON has also seen trials and initial deployments byseveral large Telco's, but these are largely used as a basis fortransmitting Ethernet via encapsulation within GEM frames (GPONEncapsulation Method) 15926 (FIG. 159). GPON was designed with verystrict timing requirements. Both EPON and GPON therefore use TimeDivision Multiple Access (TDMA), informally known as “time-sharing.”Time is divided into slots, of either fixed or variable length or longenough to contain one or more data frames (usually around 100-1000msec). During a given slot, one ONU 3002 is permitted to transmit andall others must have turned off their lasers. The OLT 3010 isresponsible for determining a transmission schedule and sending that tothe ONUs 3002 (this is sometimes considered to be a form of batchpolling by the OLT) and the ONUs must maintain an accurate clock whichis synchronized to that of the OLT in order to transmit at exactly theright time.

The number of time slots allocated to each ONU 3002 need not remainfixed. Both EPON and GPON provide flexible mechanisms to allow the OLT3010 to dynamically allocate bandwidth to ONUs according to demand andthe network operator's policy. These mechanisms are nonspecific as tothe algorithms employed, particularly in the case of EPON where theextremely simple request-based protocol leaves a lot of scope forinteresting dynamic bandwidth allocation algorithms. Extending bandwidthassignments to the mmWave technology is desirable with PON technology, achannel is broadcast to all ONUs 3002, and each frame is labelled withthe address of its target ONU. That ONU 3002 will forward the frame ontoits end user's LAN through the home gateway 3006, and all other ONUswill discard the frame. This is a form of TDMA, with the OLT 3010determining its own transmission schedule and each time slot lasting theduration of a frame.

A mmWave system 3004 can also take advantage of mmWave beam forming andbeam steering technologies to ensure QoS to the home applicationsaccessed via a home gateway 3006 in the dynamically changing networkconditions. Given the current Residential Gateway (RGW) devices 3006 donot have the ability to directly and dynamically trigger or adjust theservice flow operations based on the network conditions, the hybrid ONU3002 and mmWave Remote Units (RUs) 3008 can be designed with innovativeSDN enabled beam steering mechanisms to achieve high quality userexperience with dynamic network slicing mechanisms and optimized OLT-ONU(gPON) signaling frameworks. Millimeter wave frequencies implemented bythe mmWave system 3004 are roughly defined as bands in 24, 28, 39 and 60GHz. However, such an approach is also applicable to 3.5 GHz CBRS. ThemmWave system 3004 provides much potential for use as wireless broadbandservices with beam steering under control of SDN towards theself-installed mmWave home modems. As mentioned previously, the SDN beamsteering mechanisms and dynamic network slicing mechanisms may use thosetechniques describe in U.S. patent application Ser. No. 15/664,764,which is incorporated herein by reference.

In the vOLTHA scenario, home gateways 3006 can connect to ONUs 3002 viammWave technologies within the mmWave System 3004 in the last drop (100s of meters) where wireless access points are directly connected to ONUs3002 via mmWave RUs 3008. Hybrid virtual OLT (vOLTHA) 3010 and mmWaveFixed Broadband Wireless technology through the mmWave system 3004 canprovide self-installed access opportunities to homes and businesses. Inaddition, the synchronous nature of vOLTHA based on gPON can extenditself to map to beam steering control technology formapping/distribution of ONU traffic among multiple mmWave modems 3008with support for slicing control at home networks. In this scenario, asingle PON 3012 will be seen by an Ethernet switch as a collection ofpoint-to-point links, one per Hybrid ONU 3002+mmWave Radio Unit 3008.The PON 3012 will typically connect up to 128 ONUs 3002 to each OLT3010, and hybrid ONU−RU will connect to multiple mm-wave modemsutilizing beam steering control plans. The mmWave Modems 3008 areself-installed and reduce the need for a fiber connection to thehome/apartments as well as further provide for additional statisticalgain and aggregation points at the ONU+RU at the Ethernet layer,customers served by these PONs 3012 will be on a single large Ethernet.Furthermore, if delay and cost is not a factor, the ONU+RU's areintegrated and can be treated as IP routers with load balancing andslicing capabilities, provide statistical gain and an aggregation point.

Thus, from the operator's perspective, by bridging together all of acentral office's PONs 3012 and serving ONU+RU 3002/3008 at the Ethernetlayer, customers served by these PONs 3012 will be on a single largeEthernet. Furthermore, if delay and cost is not a factor we can threatthe ONU+RUs 3002/3008 as IP routers with opportunities for loadbalancing and additional slicing capabilities. The system may also bedesigned wherein where transmit is done at higher 60 GHz band channelfrom outside to inside and a and lower 60 GHz band channel from insideto outside.

The current ONUs 3002 in vOLTHA will be complemented with mmWave RUs3008 which will perform beam steering functions with modems installed ateach home. In practical scenarios, small cells deployed with each ONU3002 in urban outdoor environments are regularly affected by trees andpassing objects. In millimeter wave beamforming systems, theenvironmental issues such as wind-induced movement, blockage by trees,may be resolved by beam steering technologies under control of SDN whereeach wavelength uses a very narrow beam pattern. The practicalimpairments of a lamppost deployment scenario need be incorporated intothe beamforming system and system design.

Almost all modern PONs 3012 run on Ethernet at some level either used asthe native protocol on an EPON, or encapsulated in GEM on a GPON, withphysical and logical topology of a simple Ethernet PON deployment shownas follows. Ethernet is now predominantly used as a basis for the datalink layer and Internet Protocol (IP) as ubiquitous network layerprotocol. Some networks still use separate fibers for transmission ineach direction (1310 nm and 1490 nm—for bidirectional use). The EthernetPHY is responsible for providing a serialized bit stream facility (only)to the Medium Access Control (MAC) layer. The MAC is responsible fordividing the bit stream into frames. Frames are labelled with a headercontaining, source and destination MAC addresses. This enables thestatistical multiplexing of multiple hosts' frames on a single link.

FIG. 31 illustrates the interface between the ONU 3002 and the pluralityof home gateways 3006. A single optical fiber pair 3102 provides data toand from the ONU 3002. The ONU 3002 interfaces with a millimeter waveremote unit 3008, having the ability to generate RF beams 3104 that maybe directed toward one or more millimeter wave radio units 3008Bassociated with a home or business. The interface between the millimeterwave remote units 3008A and 3008B may include one or more of thebuilding penetration techniques described herein. The millimeter waveradio units 3008 provide beam steering techniques and slice controltechniques enabling the control of the transmission of databidirectionally between the ONU 3002 and home gateways 3006. Themillimeter wave remote units 3008B associated with the home or businessinterface with the home gateways 3006 to provide broadband dataconnections to the associated home or business structure.

Referring now to FIGS. 32 and 166, there are more particularlyillustrated embodiments for broadband data communications between an OLT3010 and devices located within a structure. With respect to FIG. 32,the OLT 3010 is located at a central office/MEC 3202 that may be part ofa virtual base band unit (VBBU). The OLT 3010 schedules transmissionsover the fiber 3204 to the ONU's 3002. The OLT 3010 connects to a numberof ONU's 3002 through optical fiber pairs 3204. The ONU 3002 maintainsan accurate clock to sync with the OLT 3010. Associated with the ONU3002 is a remote unit 3008. The remote unit 3008 is part of themillimeter wave system 3004 described hereinabove. The combined ONU/RVis treated as an IP router providing load-balancing and slicing andfurther providing statistical gain for signal transmission and acts asan aggregation point for received data. The combined ONV/RV alsoprovides for wireless communications with remote units associated withstructures. The remote unit 3008 is located on a light pole or towerlocated near a structure and provides the wireless last drop connectionto a home or business that replaces fiber DSL and cable.

The remote unit 3008 utilizes controlled beamforming and slice controltechniques to generate radio beams 3206 that are transmitted to anexterior millimeter wave transceiver 3208 located on an exterior of thestructure. The exterior millimeter wave transceiver 3208 repeats signalsreceive from the exterior hub and allows the signal to penetrate throughthe glass or building. The exterior millimeter wave transceiver 3208transmits the broadband data signals through a window or wall 3202 andinternal millimeter wave transceiver 3212 using one of the abovedescribed techniques for transmitting through a wall or window. Theinterior millimeter wave transceiver 3212 also utilizes beamforming andslicing techniques as described herein to transmit wireless beams 3214within the structure to a residential gateway 3216. The residentialgateway 3216 comprises a self-installed home modem that provides aninterconnection between the broadband data received from the interiormillimeter wave transceiver 3212 and devices located within thestructure that communicate with the residential gateway 3216 via wiredor wireless connections. The OLT 3010, ONU 3002, RU 3008, millimeterwave transceivers 3208/3212 and residential gateway 3216 all include ahardware abstraction layer from vOLTHA as previously described enable aSDN to control the entire end-to-end configuration of the components toaccess the last drop connection.

FIG. 33 illustrates the same structure described with respect to FIG. 32for broadband data transmissions between the OLT 3010 and the interiormillimeter wave transceiver 3212. Rather than illustrating a connectionto a residential gateway 3216, which the system may still do, a 60 GHzwireless connection to a pair of virtual reality (VR) goggles 3302 isillustrated. A 60 GHz transceiver dongle 3304, as will be more fullydescribed herein below, is inserted into a USB port of the interiormillimeter wave transceiver 3212. This provides the ability for theinterior millimeter wave transceiver 3212 to bidirectionally communicatethrough the 60 GHz transceiver dongle 3304 with the VR goggles 3302located on the interior of the structure. The VR goggles 3302 may thenbe used wirelessly with any interior computer or with a central officewithout the need for a local computer. While FIG. 33 illustrates a 60GHz wireless link to VR goggles 3302, other types of devices may alsowirelessly connected to the 60 GHz transceiver dongle 3304 in order toenable broadband data transmissions thereto. The control of datatransmissions between the optical data transmission portions and thatthe RF data transmission portions using SDN slicing as mentionedhereinabove are applicable to each of the embodiments in FIGS. 32 and33. The OLT 3010, ONU 3002, RU 3008, millimeter wave transceivers3208/3212 and VR goggles 3302 all include a hardware abstraction layerfrom vOLTHA as previously described enable a SDN to control the entireend-to-end configuration of the components to access the last dropconnection.

Referring now to FIG. 34, there is illustrated a further embodiment of ameans for wirelessly transmitting signals between an interior of abuilding 3402 and an exterior of the building 3404 through a window orwall 3406. An external transceiver processing unit 3408 is located on anexterior side 3404 of the window or wall 3406. The transceiverprocessing unit 3408 provides transmissions to/from the externalenvironment rather than transmissions from the interior of the buildingusing for example the techniques described herein above and providesprocessing of the received signals before they are transmitted to thebuilding interior. Similarly, an internal transceiver processing unit3410 is located on the interior side 3402 of the window or wall 3406.The transceiver processing unit 3410 provides transmissions to/from theinterior environment of the building rather than transmissions from theexterior of the building using for example the techniques describedherein above and provides processing of the received interior signalbefore they are transmitted to the building exterior. Inserted within aUSB interface 3412 of each of the internal transceiver processing unit3410 and external transceiver processing unit 3408 is a transceiverdongle 3414. A dongle 3414 as described herein comprises a hardwareinterface for wirelessly communicating signals between an interior andan exterior of a structure and/or other functionalities. The dongle 3414may comprise a circuit, peripheral device, etc. that connects with aprocessor on the interior and the exterior of the structure to enablecommunication between the interior and the exterior of the structureand/or the other functionalities.

The transceiver dongle 3414A located with the external transceiverprocessor 3408 is responsible for the transmission and reception ofsignals to/from the interior 3402 of the building through the window orwall 3406. The transceiver dongle 3414B located with the internaltransceiver processor 3410 transmits and receives signals to/from theexterior 3404 of the building through the window or wall 3406. Thetransceiver dongles 3414 enable the transmission of signals between theinterior and an exterior of the building in a similar manner to thatdescribed herein above with respect to other embodiments. Thetransceiver dongles 3414 consume approximately 2 W of power in eachdongle. The signals from the interior of the building are transmittedfrom or received at a Wi-Fi transceiver 3416. While one embodimentenvisions the use of a Wi-Fi transceiver, other transmission protocolscan be used for the transmissions within the building. This solution isapplicable to all types of access at any frequency band such as LTEbands including new NR radio; 3.5 GHz CBRS; 5 GHz WiFi; 24, 28 and 39Ghz licensed bands; and 60 GHz bands as well as 70/80 GHz E-band, etc.

Referring now to FIG. 35A, there is illustrated a top level blockdiagram of the embodiment of FIG. 34. The Peraso chipset 3510 isimplemented on a transceiver dongle as will be more fully describedherein below. The Peraso chipset 3510 is used for generating signalsthat can be transmitted through a window or wall between transceiverdongles 3414 located on the interior and exterior of a building. Thesignals are transmitted from antennas 3530 that comprise a patch antennaarray as described above and as will be more fully described hereinbelow. In one embodiment, the transceiver processor circuitry 3408/3410would be at least partially implemented using the RK 3399 processor3532. The RK 3399 processor 3532 comprises a low power, high performanceprocessor for computing, personal mobile Internet devices and othersmart device applications. The RK 3399 processor 3532 and the Perasochipset 3510 are powered by a power driver 3534. The power driver 3534may comprise any of the power circuitry configurations described hereinsuch as those described in FIGS. 92-94. The power driver 3534 maycomprise a single circuit that provides power to both the internal andexternal components of the communication system. Alternatively, each ofthe internal and external components may have separate power drivers3534 to power the components.

FIG. 35B more particularly illustrates a block diagram of the RK 3399processor 3532 and the Peraso chipset 3510. The RK 3399 processor 3532,which will be more fully described herein below includes a RANI memory3533, a USB 3.0 interface 3535 for interfacing with the Peraso chipset3510, a Linux/ARM 3537 port is used for connecting and ARM processor, aGPIO connection 3539, an Ethernet PHY connection 3541 and an RJ45connection 3543.

The Peraso chipset 3510 is implemented on a pair of processing devices3545 and 3547. Processing device 3545 includes a USB 3.0 interface 3549,an on-chip packet buffer 3551, connection circuitry 3553 including aSPI, I2C, PWM, UART and GPIO, and an interface 3555 comprising an IEEE802.11ad SC MAC/PHY. The second processing device 3547 includes a TX/RXfront-end, phase locked loop 3559, RX common 3561, TX common 3563, biascircuit 3565 and configuration and control 3567. Connected to the secondprocessing 3547 are first and second antenna arrays 3562.

Referring now to FIG. 35C, there is illustrated a functional blockdiagram of a further implementation of a transceiver dongle that may beused for transmitting signals through a window or wall of a structure asdescribed hereinabove. The transceivers may be implemented in the formof a dongle 3502 that may be inserted for example into a USB port of atransceiver processor located internally or externally of a building forimplementing the transmission through the window or wall. However, othertypes of connection ports may also be utilized. The dongle 3502 wouldinclude transceiver circuitry 3504, BBU circuitry 3506 and a patchantenna array 3508. The transceiver circuitry 3504 in combination withthe BBU circuitry 3506 is responsible for processing received datasignals for transmission and processing received data signals into theirindividual signals as has been described hereinabove in a number ofplaces with respect to the transmission of data using OAM or other typesof signal processing.

The BBU 3506 implements PRS 4601 WiGig baseband that is compliant withIEEE 802.11ad. The BBU 3506 includes a USB 2.0 and 3.0 device/hostsystem interface 3512 supporting link speeds of up to 2.0 Gb/s, but itis possible to configure the PRS 4601 as a multi-gigabit WiGig. The BBU3506 can modulate/demodulate all control and carrier signals up to16-QAM WiGig coding schemes (MCS0 to MCS12) up to a maximum rate of 4.62Gb/s. The BBU 3506 also includes programmable IO subsystem 3514consisting of GPIO, UART, SPI, TWI, PWM and JTAG.

The transceiver circuitry 3504 implements PRS 1126 which is a highlyintegrated, low power, single-chip mm-wave radio transceiver compliantwith the IEEE 802.11ad specification. The high performance allows thedongle 3502 to support all WiGig protocol application layers. The patchantenna array 3508 may in one embodiment comprises a PRA613-A1 which isa 4×4 concentric patch antenna in a single level array. The patchantenna array 3508 is a high efficiency, high bandwidth device with gaingreater than 8 dB across all four channels. As a stand-alone antenna, itenables independent product design when used with Peraso radiotransceiver PRS 1126. In further embodiments, the patch antenna array3508 may utilize the multilevel arrays described herein for thetransmission of data. The patch antennas making up the patch antennaarray 3508 operate in the unlicensed 57 to 66 GHz frequency band. Thepatch antennas have a gain of 8.5 dBi with plus/minus 0.5 dB gainvariation over the entire frequency band. The overall size of the patchantennas are 7.5 mm×6.5 mm×0.95 mm. The patch antenna array 3508generates beams in the H-plane having a beam width of 95° plus or minus5° and in the E-plane of 90° plus or minus 10°. The patch antennas aredesigned to work with an amplifier dish or lens and as a stand-aloneantenna.

The BBU 3506, TRX 3504 and Patch Antenna Array 3508 of are eachimplemented using the Peraso chip set 3510. As described previously,FIG. 84A illustrates a top level block diagram of a Peraso transceiverthat may be used for transmissions as described hereinabove on thetransceiver dongle. FIGS. 84B and 84C as described previously provide amore detailed application diagram of the Peraso chipset implemented onthe transceiver dongle.

FIG. 35D illustrates a block diagram of the RK 3399 processor 3532. TheRK 3399 processor 3532 integrates dual core Coretex-A72 3540 andquad-core Cortex-A53 3542 with separate NEON coprocessor within a dualcluster core 3544. The RK 3399 processor 3532 also integrates a MaliT860 MP4 GPU (graphics processing unit) 3546 within a multi-mediaprocessor 3548. The RK 3399 processor 3532 includes a good Linux supportincluding U-Boot, kernel, graphics 3550, video decoder 3552 and encoder3554 within the multi-media processor 3548. The RK 3399 processor 3532includes a CPU 3556 having a dual-core ARM Cortex-A72 MPCore processor12340 and a Quad-core ARM Cortex-A53 MPCore processor 3542. Bothprocessors are high-performance, low-power and cached applicationprocessor. The two CPU clusters comprise big clusters with the dual-coreCortex A72 3540 being optimized for high-performance and little clusterquad-core Cortex-A53 3542 being optimized for low power. The CPU 3556provides full implementation of the ARM architecture v8-A instructionset. An ARM Neon Advanced SIMD (single instruction, multiple data)provides support for accelerating media and signal processing. CCI500ensures the memory coherency between the two clusters 3540 and 3542.Each Cortex-A72 3540 integrates 48 KB L1 instruction cache and 32 KB L1data cache with 4-way set associative. Each Cortex A53 3542 integrates32 KB L1 instruction cache and 32 kB data cache separately with 4-wayset associative. The CPU 3556 further includes a 1 MB unified L2 cache3558 for the big cluster 3540 and a 512 KB unified L2 cache 3560 for thelittle cluster 3542. The CPU 3556 further provides Trustzone technologysupport.

The multi-media processor 3548 comprises an ARM Mali-T860MP4 GPU 3546that supports OpenGL ES1.1/2.0/3.0, OpenCL1.2, DirectX11.1 etc. The GPUof the multi-media processor 3548 further comprises embedded 4 shadercores with shared hierarchical tiler.

The systems memory comprises external memory interface 3562 and embeddedmemory components 3564. The external memory interface 3562 includes adynamic memory interface (DDR3/DDR3L/LPDDR3/LPDDR4) 3566 that iscompatible with JEDEC standard DDR3-1866/DDR3L-1866/LPDDR3-1866/LPDDR4SDRAM. The dynamic memory interface 3566 supports two channels, whereineach channel has 16 or 32 bits data width. The dynamic memory interfacealso supports up to two ranks (chip selects) for each channel totaling 4GB(max) address space. Maximum address space of one rank in a channel isalso 4 GB, which is software-configurable. The eMMC Interface 3568 isfully compliant with JEDEC eMMC 5.1 and eMMC 5.0 specification. Theinterface 3568 supports HS400, HS200, DDR50 and legacy operating modes.SD/MMC Interface 3570 includes two MMC interfaces which can beconfigured as SD/MMC or SDIO. The SD/MMC interface 3570 is compatiblewith SD3.0, MMC ver4.51.

System peripherals 3572 include but are not limited to timers 3574including 14 on-chip 64 bits timers in SoC with interrupt-basedoperation for non-secure application and 12 on-chip 64 bits Timers inSoC with interrupt-based operation for secure applications. PWM 3576include four on-chip PWMs with interrupt-based operation. A WatchDog3578 includes three Watchdogs in SoC with 32 bits counter widths.

The multi-media processor 3548 comprises an ARM Mali-T860MP4 GPU 3546that supports OpenGL ES1.1/2.0/3.0, OpenCL1.2, DirectX11.1 etc. The GPUof the multi-media processor 3548 further comprises embedded 4 shadercores with shared hierarchical tiler.

Video components of the RK 3399 processor 3532 include real-time videodecoder of MPEG-1, MPEG-2, MPEG-4, H.263, H.264, H.265, VC-1, VP9, VP8,MVC; H.264 10 bit up to HP level 5.1: 2160p@60 fps (4096×2304); VP9:2160p@60 fps (4096×2304); H.265/HEVC 10 bit: 2160p@60 fps (4096×2304);MPEG-4 up to ASP level 5: 1080p@60 fps (1920×1088); MPEG-2 up to MP:1080p@60 fps (1920×1088); MPEG-1 up to MP: 1080p@60 fps (1920×1088);H.263: 576p@60 fps (720×576); VC-1 up to AP level 3: 1080p@30 fps(1920×1088); VP8: 1080p@60 fps (1920×1088); MVC: 1080p@60 fps(1920×1088); support video encoders for H.264, MVC and VP8.

The system display components include embedded two VOP, outputs from thefollowing display interface: one or two MIPI-CSI port 3580, one eDP port3581, one DPI port 3582, and one HDMI port 3583. The ports support AFBCfunction co-operation with the GPU. The HDMI interface 3584 comprises asingle physical layer PHY with support for HDMI 1.4 and 2.0 operation aswell as support HDCP 1.4/2.2. The MIPI interface includes embedded 3MIPI PHY, MIPI0 only for DSI, MIPI1 for DSI or CSI, MIPI2 only for CSI.Each port has 4 data lanes that provide up to 6.0 Gbps data rate. TheeDP interface is compliant with eDPTM specification, version 1.3 for upto 4 physical lanes of 2.7/1.62 Gbps/lane. A display port interface iscompliant with display port specification, version 1.2 and is compliantwith HDCP2.2 (and back compatible with HDCP1.3). There is only onedisplay port controller built-in RK 3399 which is shared by 2 Type-C.

Connectivity components 3585 include a camera interface and imageprocessor that include one or two MIPI-CSI input interfaces and twoembedded ISP (Image Sensor Processors). A maximum input resolution ofone ISP is 13M pixels. Connectivity components include an embedded 2Type-C PHY 3586. The connectivity components 3585 are compliant with USBType-C Specification, revision 1.1 and with USB Power DeliverySpecification, revision 2.0. Connection components 3585 haveattach/detach detection and signaling as DFP, UFP and DRP as well asplug orientation/cable twist detection. The connections support USB3.0Type-C and DisplayPort 1.2 Alt Mode on USB Type-C, two PMA TX-only lanesand two PMA half-duplex TX/RX lanes (can be configured as TX-only orRX-only). The connectivity components provide up to 5 Gbps data rate forUSB3.0, up to 5.4 Gbps (HBR2) data rate for DP1.2, can support 1/2/4lane modes.

Audio components of the RK 3399 processor 3532 include three I2S/PCM inSoC 3587. I2S0/I2S2 supports up to eight channels TX and eight channelsRX. I2S1 supports up to two channels TX and two channels RX. I2S2 isconnected to an HDMI and DisplayPort internally. I2S0 and I2S1 areexposed for peripherals. Audio components further include SPDIF 3588that supports two 16-bit audio data store together in one 32-bit widelocation. SPDIF 3588 also supports bi-phase format stereo audio dataoutput and 16 to 31 bit audio data that is left or right justified in32-bit wide sample data buffer. Finally, SPDIF 3588 supports 16, 20, 24bits audio data transfer in a linear PCM mode.

Connectivity 3585 further includes an SDIO interface 3589 that iscompatible with SDIO 3.0 protocol. A GMAC 10/100/1000M Ethernetcontroller supports 10/100/1000-Mbps data transfer rates with the RGMIIinterfaces and supports 10/100-Mbps data transfer rates with the RMIIinterfaces. A SPI controller 3590 includes six on-chip SPI controllers.A UART Controller 3591 includes five on-chip UART controllers. A I2Ccontroller includes nine on-chip I2C controllers.

Connectivity components 3585 further include two embedded USB 2.0 Hostinterfaces 3594, two embedded USB OTG3.0 interfaces and one PCIe portcompatible with PCI Express Base Specification Revision 2.1.

Other RK 3399 components include an embedded two channel TS-ADCtemperature sensor 3595, 6-channel single-ended 10-bit successiveapproximation register analog-to-digital converter (SAR-ADC) 3596 thatprovides a conversion speed range up to 1 MS/s sampling rate and two1024 bits (32×32) high-density electrical fuses (eFuse) that areintegrated.

Referring now to FIG. 36 there is illustrated a block diagram of themodulation system. Signals to be transmitted are provided at input 3674in a digital format and converted from digital to analog format at thedigital to analog converter 3676 responsive to a clock signal from clockgenerator 3670. The analog signal is modulated within modulator 3678responsive to the analog signal and control signals from the phaselocked loop/local oscillator block 3666. The modulated signals aretransmitted from antenna 3662B in one of the configurations describedhereinbelow from the Peraso transceiver 3660. The Peraso chipset is moreparticularly described in the Peraso W110 WiGig Chipset Product Briefdated Dec. 18, 2015 which is incorporated herein by reference.

Referring now to FIGS. 37A and 37B, there is illustrated a more detailedapplication diagram of the Peraso chipset. While the Peraso chipset inthe 60 GHz band has been described, it will be realized by one skilledin the art that the chipset may utilize any frequency where the repeaterenables extension of signal transmission capabilities. Examples include,but are not limited to, millimeter bands, 28 GHz band, 39 GHz band, 2.5GHz band, CBRS band (3.5 GHz) and Wi-Fi band (5 GHz). The Peraso chipsetcomprises the W110 chipset that is targeted for use with WiGigapplications. The Peraso chipset employs a PRS 1125 integrated circuit3702 and PRS 4001 integrated circuit 3704 to implement the IEEE 802.11adfunctionality. The Peraso chipset implements a complete superspeed USB3.0 to WiGig solution. The PRS 4001 low power WiGig baseband integratedcircuit 3702 incorporates the analog front end 3706 including digital toanalogue converters 3708, analog-to-digital converters 3710 and a phaselocked loop 3712. The PRS 4001 circuit 3702 further includes thebaseband physical layer 3714, Mac layer 3716 and two RISC CPU cores. ThePRS 4001 circuit 3702 is IEEE 802.11ad compliant. A USB 2.0 and 3.0interfaces 3724 enable USB communications. The PRS 4001 circuit 3702supports seamless connection to all Peraso radios.

The PRS 1125 integrated circuit 3704 is a single chip direct conversionRF transceiver providing 60 GHz single ended receiver and transmitinterfaces. The PRS 1125 circuit 3704 provides a transmit output powerof up to 14 dBm, better than −21 dB transmit EVM (16-QAM), receivernoise less than 5 dB and a receiver conversion gain of greater than 70dB. Integrated single ended 60 GHz antenna interfaces include a transmitdata path 3718 and a received data path 3720. A phase locked loop 3722tunes to all channels of IEEE 802.11ad using an integrated controller.The Peraso chipset provides for wireless storage, wireless display andmulti-gigabyte mobile wireless applications. The antennas 3726 compriseNA graded patch antennas with 8.5 dBi gain across the entire 60 GHzband.

Full-duplex communications between Peraso chipset transceivers may becarried out in a number of fashions in order to control throughputtherebetween. As illustrated in FIG. 38, communications between thefirst Peraso transceiver 8502 and a second Peraso transceiver 8504 maybe carried out in series over a single communications channel 3806. Inthis case, the data is transmitted serially one item after the otherover the single communications channel 3806. FIG. 37 illustrates aparallel full-duplex transmission configuration. In this configuration,transmissions between transceiver 3702 and transceiver 3704 occur overmultiple channels 3708 operating in parallel. In this configuration,different data streams may be transmitted at the same time over theparallel communication channels 3708 in order to increase datathroughput. In the parallel configuration, a data stream is petitionedin two multiple sub-streams and sent on the separate parallel channels3708. The results may then be combined together at the receiver 3704.

Communications between Peraso chipset transceivers may be carried out ina number of fashions in order to control throughput therebetween. Asillustrated in FIG. 38, communications between the first Perasotransceiver 8502 and a second Peraso transceiver 8504 may be carried outin series over a single full-duplex communications channel 3806. In thiscase, the data is transmitted serially one item after the other over thesingle communications channel 3806. FIG. 39 illustrates a paralleltransmission configuration. In this configuration, full-duplextransmissions between transceiver 3902 and transceiver 3904 occur overmultiple channels 3908 operating in parallel. In this configuration,different data streams may be transmitted at the same time over theparallel communication channels 3908 in order to increase datathroughput. In the parallel configuration, a data stream is partitionedinto multiple sub streams and sent on the separate parallel channels3908. The results may then be combined together at the receiver 3904.

FIG. 40 more particularly illustrates a transceiver dongle 4002 and themultilevel patch antenna array 4004 for transmitting and receiving OAMsignals. The transceiver dongle 4002 interfaces with other devices usinga USB connector 4006. The transceiver dongle 4002 also includes thepatch antenna array 4004 which includes a first layer of patch antennas4008 in a circular array. The patch antennas 4008 would provide for thetransmission of signals across a window or wall to a second transceiverdongle. In an alternative embodiment, a second layer of patch antennasin a circular array within the first array of patch antennas may becombined with the first layer of patch antennas to enable thetransmission of multiple signals. The first layer of patch antennas 4008would transmit, for example, signals having an OAM function including anl=+1 helical beam and the second layer of patch antennas would receivesignals having an OAM function including an l=−1 helical beam. Each ofthe first layer patch antennas 4008 and the second layer of patchantennas are at different level layers as described herein above withrespect to FIGS. 105-114 to enable the transmission and reception of OAMsignals between the interior and the exterior of the building.

FIG. 41 illustrates a side view of the transceiver dongle 4102. The sideview illustrates the USB connector 4104 that is used for interconnectingthe transceiver dongle 4102 with the processing units described withrespect to FIG. 170. A baseband IC 4106 up/down converts signals betweenbaseband levels and RF levels. The radio IC 4108 transmits and receivesthe RF signals received by antenna 4110. The antenna 4110 in a preferredembodiment comprises the multilevel patch antenna described hereinabove.

The operation of the transceiver dongles have been tested at variousdistances. When two transceiver dongles are spaced at a distance of 25cm (approximately 10 inches), the transceiver dongles have beendetermined to have a throughput of approximately 1.53 Gb per secondwithout glass in the open air and with a throughput of 734 Mb per secondthrough glass. When two transceiver dongles are spaced at a distance of15 cm (approximately 6 inches), the transceiver dongles have beendetermined to have a throughput of approximately 1.29 Gb per secondwithout glass in the open air. When the transceiver dongles are placedupon opposite sides of window glass at a distance of approximately 2 cm(1 inch), the throughput is 1.5 Gb per second.

FIG. 42 more particularly illustrates a transceiver dongle 4202 and themultilevel patch antenna array 4204 for transmitting and receiving OAMsignals. The transceiver dongle 4202 interfaces with other devices usinga USB connector 4206. The transceiver dongle 4202 also includes thepatch antenna array 13404 which includes a first layer of patch antennas4208 in a circular array and a second level of patch antennas 4210within a circular array. The first layer of patch antennas 4208 wouldtransmit, for example, signals having an OAM function including an l=+1helical beam and the second layer of patch antennas 4210 would receivesignals having an OAM function including an l=−1 helical beam. Each ofthe first layer patch antennas 4208 and the second layer of patchantennas 4210 are at different level layers as described herein above toenable the full-duplex transmissions using OAM signals.

Referring now to FIG. 43, there is illustrated the three level patchantenna array 4302 including a first level of a substantially circularpatch antenna array 4304, a second level of a substantially circularpatch antenna array 4306 and a third level of a substantially circularpatch antenna array 4308. Each of the levels of patch antenna arrayscomprise a plurality of patch antennas 4310 that are located on asubstrate 4312. Each of the levels include a separate input or outputdepending on whether the array comprises a transmitting or receivingarray. Antenna array 4304 comprises a transmit array having an input4314 for receiving a signal to be transmitted by the array 4304. Antennaarray 4306 comprises a receive patch antenna array having an output 4316for outputting a received signal received by the patch antenna array.Antenna array 4308 also comprises a transmit array having an input 4318for receiving a signal to be transmitted by the array 4308. Bytransmitting the same signals from the patch antenna array 4304 andpatch antenna array 4308, destructive interference will enablecancellation of any of the transmitted signals received by the receivepatch antenna array 4306 as will be more fully described hereinbelow.The improvement of signal interference between the transmitted andreceived signals may also be improved by the selection of a substrate4312 for containing the patch antennas 4310 that has characteristics forlimiting signal interference between antenna layers.

Referring now to FIG. 44 there is illustrated a top view of themultilevel patch antenna array 4402 comprising the bottom layer 4404,the mid-layer 4406 and the top layer 4408. The bottom layer 4404includes a first circular array of patch antennas 4410. The mid-layer4406 includes a second circular array of patch antennas 4412. The toplayer 4408 includes a third circular array of patch antennas 4414. Eachof the layers are concentric with the mid-layer 4406 and top layer 4408being within the area of the bottom layer 4404, and the top layer 4408being within the area of the mid-layer 4406. This enables unimpededtransmission and reception of signals by the associated patch antennasin each layer.

Referring now also to FIG. 45, there is provided a top-level view morefully illustrating the size of the bottom layer 4404, mid-layer 4406 andtop layer 4408. The distance between the edge of the top layer 4408 andthe mid-layer 4406 will have an established value equal to d. Thedistance between the edge of the mid-layer 4406 and the bottom layer4404 is defined in accordance with the distance d to have a distance ofdλ/2. This configuration of the distances between bottom layer 4404,mid-layer 4406 and top layer 4408 causes the signal from the twotransmit layers on the bottom layer 4404 and the top layer 4408 to adddestructively causing significant attenuation in the signal received atthe receive antenna on the mid-layer 4406 from the bottom layer and thetop layer.

FIG. 46 provides an illustration of the analog and digital cancellationprocess provided by the processing circuitry of a full-duplex system.Within the in-band full-duplex terminal 4602, there is an input forreceiving the transmit bits 4604. The transmit bits 4604 are applied tocoding and modulation circuitry 4606 within the digital domain 4608 toapply digital coding and modulation to the transmit bits 4604. The codedand modulated bits are passed on to N transmit chains 4610 for furtherprocessing. Each of the N transmit chains 4610 include adigital-to-analog (DAC) converter 4612 for converting the signal fromthe digital domain to the analog domain. The analog signal from the DAC4612 is applied to one input of a mixer circuit 4614 that is mixed witha signal from an oscillator 4616 for up-conversion. The up-convertedsignal from the mixer circuit 4614 is applied to a high power amplifier4618 for amplification. The amplified signal is transmitted from anassociated antenna 4620 and to a canceller circuit 4622. The transmittedsignals 4624 may be reflected from nearby scatterers 4626 as a reflectedpath signal 4628 to the receive antenna 4630. The transmitted signal4624 may also create a direct path signal 4632 to the receive antenna4630. The direct path signal 4632 and that the reflected path signals4628 comprise the combined total self-interference 4634 that interfereswith the desired received signal 4636 at the receive antennas 4630. Thetransmit antenna 4620 and receive antennas 4630 comprise part of thepropagation domain 4635. The total self-interference 4634 may becanceled from the signals received at the receive antennas 4630 usingthe canceller circuit 4622.

The canceller circuit 4622 generates a cancellation signal that isapplied at an adder circuit 4638 on each of the N receive chains 4640 toremove the total interference signal 4634 from the received signal 4636.The canceller circuit 4622 generates the cancellation signal for theadder circuit 4638 responsive to cancellation control signals 4642applied from the digital domain 4608, and the transmit signals 4624 fromeach of the N transmit chains 4610. Each of the N receive chains 4640include a low noise amplifier (LNA) 4644 for amplifying the receivedsignal that has analog cancellation applied thereto. The output of theLNA 4644 is applied to a mixing circuit 4646 along with an oscillationsignal from oscillator 4648 to down-convert the receive signal. Thecanceller circuit 4622, adders 4638, LNA 4644, mixer circuit 4646 andoscillator 4648 are all part of the analog circuit domain 4650.

The down converted signal from the mixer circuit 4646 is applied to aninput of an analog to digital converter (ADC) 4652 to convert the signalfrom the analog domain to the digital domain in each of the N receivechains 4640. The digital received signals have digital interferencecancellation, demodulation and decoding applied to them within thedigital domain 4608. The processed signals are output as receive bits4654.

Referring now to FIG. 47, there is illustrated a block diagram of theanalog and digital cancellation circuitry. The digital cancellationcircuitry 4702 limits all linear and nonlinear distortion within thereceive signal responsive to digital inputs of the transmitted bits Tb4704. The transmitted bits 4704 are provided to a digital to analogconverter 4706 followed by a mixing circuit 4708 before being amplifiedby a power amplifier 4710 on the transmitter side of the circuit. Thetransmitted signal from the output of the power amplifier 4710 isapplied as an input to analog cancellation circuit 4712 and to a firstinput of the circulator 4714 that applies the transmitted signal to anantenna 4716. The antenna 4716 also receives signals that are providedto the circulator 47144 output at a third port of the circulator thatare received by the antenna 4716 at a second port. The receive signalincludes the desired received signal and any interference caused by thetransmitted signals and reflected transmitted signals (R+aT). Thereceive signal is applied to the receiver circuitry including asummation circuit 4718 that receives a cancellation signal from theanalog cancellation circuit 4712. The summation circuit 4718 cancels theanalog interference portion of the received signal that is applied to alow noise amplifier 4720. The amplified received signal from the lownoise amplifier 4720 is applied to a mixing circuit 4722 for downconversion. The down-converted signal is applied to the digitalconverter 4724 for conversion from the analog-to-digital domain. Thedigital signal is applied to a summation circuit 4726 for combinationwith a digital cancellation signal received from the digitalcancellation circuitry 4702. This process removes linear and nonlineardistortion from the digital signal to output the received signal 4726.

The analog cancellation circuit 4712 receives an input from the outputof the power amplifier 4710 of the transmitter. From the signal, a fixeddelay d_(N) is determined at 4728 for each transmission chain 4610.Next, the analog cancellation circuit 4712 determines a variableattenuator a_(N) 4730 for each transmission chain 4610 responsive to thesignal from the power amplifier 4710 processed by the fixed delays 4728and a control input from control algorithm 4732. The outputs from eachof the variable attenuators 4730 are summed at a summation circuit 4734before being output to the summation circuit 4718 of the receiver.

Referring now to FIG. 48, there is provided a functional illustration ofthe transmit and receive signal paths for the analog and digitalcancellation process. Beginning with the transfer, the encoder 4802provides the transmitted signal to a digital to analog converter (DAC)4804 and to digital interference cancellation circuitry 4806. The signalprovided to the digital interference cancellation circuit 4806 isreferred to as a digital interference reference signal and is used forcancellation. The analog converted signal from the DAC 4804 is providedto a baseband to RF conversion circuit 4810 and to RF analogtransmission circuitry 4812. The transmitted signal passes through afirst power splitter 4814 that splits the signal for transmission overantenna TX2 4806 or to a second power splitter 4818 that splits thetransmitted signal for provision to antenna TX1 4820 and to RFinterference cancellation 4822.

The receive signal path receives signals at receiving antenna RX 4824.In order to improve destructive interference cancellation by theantennas. The receive antenna RX 4824 is spaced from antenna TX1 4820 adistance d and is spaced from antenna TX2 4816 a distance d+λ2. Theplacement of the transmit and receive antennas in the manner describedreduces self-interference based upon antenna cancellation. The antennacancellation evaluates limits with respect to bandwidth of the signalbeing transmitted and the sensitivity of antenna cancellation toengineering errors. Antenna cancellation can potentially achieve 20 dBreduction in self-interference. The effects of using two transmitantennas for antenna cancellation can be postulated as follows. If thewavelength of transmission is λ, and the distance of the receive antenna4824 is d from one transmit antenna 4820, the other transmit antenna4816 is placed at d+λ2 away from the receive antenna. This causes thesignal from the two transmit antennas to add destructively, thus causingsignificant attenuation in the signal received, at the receive antenna.

The receive signal is input to the RF interference cancellation 4822 toan OHx220 cancellor 4826. Also input to the OHx220 cancellor 4826 is thetransmitted signal provided to antenna TX 14807. The signal is referredto as the RF interference reference signal. The OHx220 cancellor 4826 isoutput to an RF analog receiver 4828. The signal is next forwarded forRF to baseband down conversion at RF to baseband down converter 4830.The down converted signal is provided to analog to digital converter4832 before being provided to the digital interference cancellation 4806for digital signal cancellations responsive to the previously discusseddigital interference reference signal. The receive signal is decoded bythe decoder 4834.

With respect to the digital interference cancellation circuit 4806,there is extensive existing work that describes various digitalcancellation techniques. Traditionally, digital cancellation is used bya receiver to extract a packet from a desired transmitter after thepacket has collided with a packet from an unwanted transmitter. To dothis, the receiver first decodes the unwanted packet, re-modulates itand then subtracts the packet from the originally received collidedsignal. In case of canceling self-interference for full-duplex, thetransmitted symbols are already known, and thus decoding is notnecessary in order to reconstruct a clean signal. Instead of decoding,coherent detection is used to detect the self-interfering signal. Thedetector correlates the incoming received signal with the cleantransmitted signal, which is available at the output of the transmitter.The main challenge in subtracting the known signal is in estimating thedelay and phase shift between the transmitted and the received signals.As the detector has the complete knowledge of originally transmittedsignal, the detector uses this signal to correlate with the incomingsignal to detect where the correlation peaks occur. The correlation peaktechnique gives both the delay and the phase shift needed to subtractthe known signal from the received signal. Thus, this technique, unlikesome of the digital interference techniques, does not require anyspecial preamble or postamble and is backwards compatible. Moreover,this technique is modulation-independent as long as the clean signal canbe constructed from the transmitted signal. Coherent detection candetect the self-interference signal even when the self-interferencesignal is weaker than the received signal. Therefore, digitalinterference cancellation can improve the SNR level even when thereceived signal is stronger than self-interference. This property isuseful when operating with variable data rates to allow using higherdata rates for high SNR links.

Typical interference cancellation also requires compensating for clockdrift between the transmitter and receiver. Since the transmitter andreceiver daughterboards in a full-duplex node share the same clock,there is no clock drift. However, since the daughterboards use separatePLL logic, there can be a jitter introduced into the transmissions.

Full Duplex

FIG. 49 illustrates a simplified block diagram of an RF receiver. Thereceiver antenna 4904 receives the RF signal 4906. The RF signal 4906 isprovided to an RF mixer 4908 where the signal is mixed with a carrierfrequency 4910 to generate the baseband signal 4912. The baseband signalis provided to an analog to digital converter 4914 to generate basebandsignal 4916. A baseband demapper 4918 demaps the baseband signal 4916into the received bits 4920. Interference between the received RF signal4906 and an RF signal from a transmitting antenna can interfere eachother causing distortion of the received bits 4920. Thus, the ability toovercome this interference using full-duplex transmission techniques canimprove signal reception. As discussed above, existing techniques ofovercoming interference in full-duplex systems have a variety oflimitations.

Motivated by these limitations, recent work has proposed antennaplacement techniques. The state of the art in full duplex operates onnarrowband 5 MHz signals with a transmit power of 0 dBm (1 mW). Thedesign achieves this result by augmenting the digital and analogcancellation schemes described above with a novel form of cancellationcalled “antenna” cancellation as shown in FIG. 50. The separationbetween the receive antenna 5002 and the transmit antennas 5004attenuates the self-interference signal, but the separation is notenough. A second transmit antenna 5006 placed in such a way that the twotransmit signals interfere destructively at the receive antenna. This isachieved by having one-half wavelength distance offset between the twotransmit antennas. The receive antenna 5010 utilizes RF interferencecancellation 5008 to attempt to overcome the transmission signalinterference and processes the signal using analog to digital conversionat ADC 5010 and further digital cancellation techniques at digitalcanceler 5012. This design thus uses multiple cancellation techniquesincluding the antenna cancellation, RF interference cancellation anddigital cancellation.

This design still has limitations. The first limitation relates to thebandwidth of the transmitted signal. Only the signal at the centerfrequency is perfectly inverted in phase at the receiver 5002 so it isfully cancelled. However, the further away a signal is from the centerfrequency, the further the signal shifts away from perfect inversion anddoes not cancel completely. Cancellation performance also degrades asthe bandwidth of the signal to cancel increases.

The cancellation is highly frequency selective and modulation approachessuch as OFDM which break a bandwidth into many smaller parallel channelswill perform even more poorly. Due to frequency selectivity, differentsubcarriers will experience drastically different self-interference.Another limitation is the need for three antennas. Full duplex can atmost double throughput, but a 3×3 MIMO array can theoretically triplethroughput which suggests that it may be better to use MIMO. The thirdlimitation is that the full duplex radio requires manually tuning thephase and amplitude of the second transmit antenna to maximizecancellation at the receive antenna.

FIG. 51 illustrates a block diagram of a full-duplex system. A fullduplex system radio can be created that requires only two antennas 5102,has no bandwidth constraint, and automatically tunes itsself-interference cancellation. To achieve this, a radio needs to havethe perfect inverse of a signal so that it can be fully cancelled out. Abalun transformer 5104 can be used to obtain the inverse of aself-interference signal then use the inverted signal to cancel theinterference. This technique is called balun passive cancellation anduses high precision passive components to realize the variableattenuation and delay in the cancellation path.

There are practical limitations to this technique, for example, thetransmitted signal on the air experiences attenuation and delay. Toobtain perfect cancellation the radio must apply identical attenuationand delay to the inverted signal, which may be hard to achieve inpractice. The balun transformer 5104 may also have engineeringimperfections such as leakage or a non-flat frequency response.

Referring now also to FIG. 52, the balun transformer 5104 splits thetransmit signal and uses wires of the same length for theself-interference path 5202 and the cancellation path 5204. The passivedelay line and attenuator provide fine-grained control to match phaseand amplitude for the interference and cancellation paths 5202, 5204 tomaximize cancellation. Balun cancellation is not perfect across theentire band, and this is because the balun circuit is not frequencyflat. Based on FIG. 53, the best possible cancellation can be obtainedwith the balun transformer 5104 and phase-offset cancellation for agiven signal bandwidth. FIG. 54 shows the best cancellation achievedusing each method.

FIG. 54 shows that if the phase and amplitude of the inverted signal areset correctly, the balun cancellation can be very effective. If one canestimate the attenuation and delay of the self-interference signal andmatch the inverse signal appropriately, then one can self-tune acancellation circuit. The auto-tuning algorithm would adjust theattenuation and delay such that the residual energy after baluncancellation would be minimized. Let g and τ be the variable attenuationand delay factors respectively, and s(t) be the signal received at theinput of the programmable delay and attenuation circuit. The delay overthe air relative to the programmable delay is τ_(a). The attenuationover the wireless channel is g_(a). The energy of the residual signalafter balun cancellation is:E=∫ _(T) _(o) (g _(a) s(t−τ _(a))−gs(t−τ))² dtwhere T_(o) is the baseband symbol duration. The goal of the algorithmis to adjust the parameters g and τ to minimize the energy of theresidual signal.

FIG. 52 shows the balun cancellation circuit, but it only handles thedominant self-interference component. A node's self-interference mayhave other multipath components which are strong enough to interferewith reception. The balun circuit may also distort the cancellationsignal slightly which introduces some leakage. A full duplex radio usesdigital cancellation to prevent the loss of packets which a half-duplexradio could receive.

The digital cancellation has three novel achievements compared toexisting software radio implementations. It is the first real-timecancellation implementation that runs in hardware. The secondachievement is that it is the first cancellation implementation that canoperate on 10 MHz signals. Finally, it is the first digital cancellationtechnique that operates on OFDM signals.

Digital cancellation has two components: estimating theself-interference channel, and using the channel estimate on the knowntransmit signal to generate digital samples to subtract from thereceived signal. The radio uses training symbols at the start of atransmitted OFDM packet to estimate the channel. Digital cancellationmodels the combination of the wireless channel and cancellationcircuitry effects together as a single self-interference channel. Due toits low complexity, the least squares algorithm is used in theestimation. The least squares algorithm estimates the channel frequencyresponse of each subcarrier:

${{\hat{H}}_{s}\lbrack k\rbrack} = {\frac{1}{M}\left\lbrack {\frac{1}{X\lbrack k\rbrack}\left( {\sum\limits_{m = 1}^{M}{Y^{(m)}\lbrack k\rbrack}} \right)} \right\rbrack}$

The radio applies the inverse fast Fourier transform to the frequencyresponse to obtain the time domain response of the channel. This methodof estimating the frequency response uses the least squares algorithm tofind the best fit that minimizes overall residual error. The radioapplies the estimated time domain channel response to the knowntransmitted baseband signal and subtracts it from the received digitalsamples. To generate these samples, the hardware convolves with the FIRfilter. The output i[n] of the filter:

${i\lbrack n\rbrack} = {\sum\limits_{k = 0}^{N - 1}{{{\overset{\hat{}}{h}}_{s}\lbrack k\rbrack}{s\left\lbrack {n - k} \right\rbrack}}}$

The radio subtracts the estimates of the transmit signal from thereceived samples r[n]:

${\overset{\hat{}}{r}\lbrack n\rbrack} = {{{r\lbrack n\rbrack} - {i\lbrack n\rbrack}} = {{\sum\limits_{k = 0}^{N - 1}{{h_{d}\lbrack k\rbrack}{d\left\lbrack {n - k} \right\rbrack}}} + {\sum\limits_{k = 0}^{N - 1}{\left( {{h_{s}\lbrack k\rbrack} - {{\overset{\hat{}}{h}}_{s}\lbrack k\rbrack}} \right){s\left\lbrack {n - k} \right\rbrack}}} + {z\lbrack n\rbrack}}}$

Where d[n] and h_(d)[n] are transmitted signal and channel impulseresponse from the intended receiver, and z[n] is additive white Gaussiannoise.

As described above, full duplex communication involves simultaneoustransmission and reception of signals over an available bandwidthbetween transmission sites. The various details of full-duplexcommunications and other full-duplex wireless transmission techniquesare more fully described in “Practical, Real-time, Full DuplexWireless,” Jain et al., MobiCom '11, Sep. 19-23, 2011, Las Vegas, Nev.,USA, 2011, which is incorporated herein by reference in its entirety.

Referring now to FIG. 55, as referenced above, a communication system5502 including a first transceiver 5504 and a second transceiver 5506communicate with each other over communication channel 5508 from thefirst transceiver to the second transceiver and communication channel5510 from the second transceiver to the first transceiver. The firstcommunication channel 5508 and the second communication channel 5510will interfere with each other if transmitted using the same frequencyor channel. Thus, some manner for overcoming the interference betweenthe channels is necessary in order to enable the transmissions from thefirst transceiver 5504 to the second transceiver 5506 to occur at a sametime. One manner for achieving this is the use of full-duplexcommunications. Some embodiments for full-duplex communication have beendescribed hereinabove.

FIG. 56 illustrates a full duplex communication system 5602 wherein afirst transceiver 5604 is an communication with the second transceiver5606. In the implementation of the communications channel fromtransceiver 5604 to transceiver 5606 has incorporated therein an orbitalangular (OAM) of +l₁, and the communication channel from transceiver5606 to transceiver 5604 has incorporated therein an OAM of −l₁. Thetransceiver 5604 transmits signals having the OAM+l₁ function appliedthereto, and the transceiver 5604 transmits signals having the OAM −l₁signal applied thereto to prevent interference therebetween. The OAMsignals each comprise orthogonal functions that are orthogonal to eachother. Since the signals are orthogonal to each other, they do notinterfere with each other even when being transmitted over the samefrequency or channel. This achieves isolation between the transmittingand the receiving channels. This allows the full-duplex communicationswith transmissions from transceiver 5604 to transceiver 5606 and fromtransceiver 5606 to transceiver 5604 to occur at the same time withoutinterfering with each other. For longer distances within opticaltransmissions systems, lenses may be used to focus the beams transmittedbetween transmitters 5604 and 5606. This enables beams to be transmittedover a further distance. The applied orthogonal functions can be orbitalangular momentum, Laguerre-Gaussian functions or others in a cylindricalcoordinate system for transmitting and receiving.

The full-duplex communications capability and potential interferenceissues are more fully illustrated with respect to FIG. 57. In FIG. 57,the transceiver 5604 includes a transmitting antenna 5702 and areceiving antenna 5704. The second transceiver 5606 consist of atransmitting antenna 5706 and receiving antenna 5708. The transmittingantenna 5702 transmits a signal having a +l₁ OAM function appliedthereto. The +l₁ signal is received by the receiving antenna 5708, bythe transmitting antenna 5706 and by the receiving antenna 5704.Similarly, the transmitting antenna 5706 transmits a signal having a −l₁OAM function applied thereto. The transmitted −l₁ OAM processed signalis received at the receiving antenna 5708, the receiving antenna 5704and the transmitting antenna 5702. In this manner, both the +l₁ signalsand the −l₁ signals are received at each antenna. By applying theorthogonal OAM functions to the transmitted signals, each of thereceivers associated with the antennas 5702-5708 may process the signalsin such a manner as to only look for transmitted signals having aparticular OAM value applied thereto. Signals having another OAM valueapplied thereto are ignored. Thus, antennas 5702 and 5706 would onlyconcentrate on transmitting the +l₁ signals and the −l₁ signals,respectively. The receiver antenna 5704 would be configured to only payattention to received −l₁ signals and the receiver antenna 5708 wouldonly pay attention to received +l₁ signals. Thus, by utilizing differentorthogonal functions, the receiver RX₁ may be configured to only processsignals having the orthogonal function −l₁ applied thereto. The receivedsignals including the orthogonal function +l₁ are ignored. The receiverRX₂ functions in a similar manner in only processes the received signalshaving the orthogonal function +l₁ applied thereto while ignoring theorthogonal function −l₁. In this manner, interference between thesimultaneously transmitting full-duplex transmit and receive channelsmay be managed. Other types of orthogonal functions other than OAM mayalso be used.

FIG. 58 illustrates a functional block diagram of a transceiver that maybe utilized for each of the transceivers 5604 and 5606 that areillustrated with respect to FIGS. 56 and 57. The transceiver 5802includes a data interface 5804 enabling the transceiver to receive oneor more data streams for transmission from the transceiver 5802 via anantenna 5806. Information received over the data interface 5804 isprocessed via an RF/optical signal modulator/demodulator 5808 thatmodulates signals to be transmitted from the transceiver 5802 using theapplicable RF or optical data transmission protocol, and fordemodulating received RF/optical signals using the applicable protocol.The OAM signal processing circuitry 5810 is used for applying theorbital angular momentum or other orthogonal function to the modulateddata signal that is to be transmitted from the antenna 5806.Additionally, the OAM signal processing circuitry 5810 may be used forremoving of the OAM or orthogonal function that is applied to receivedsignals prior to their demodulation by the demodulator 5808. Thefull-duplex processing circuitry 5812 is used for controlling thereceived signals that are to be processed by the transceiver 5802. Asdescribed earlier with respect to FIG. 10, when two signals are receivedat RX1 1004 having the OAM value +l₁ applied thereto and the otherhaving the OAM value −l₁ applied thereto, the full-duplex processingcircuitry 5812 will control which of the received signals are to beprocessed by the receiver. This will require the full-duplex processingcircuitry 5812 to identify the OAM value or orthogonal function valuethat has been applied to the receive signal in order to determinewhether the received signal should be processed. Finally, thetransmitter 5814 is used for outputting the generated signals from theantenna 5806 of the transceiver 5802.

The RF/optical modulator/demodulator 5808 and OAM signal processingcircuitry 5810 may utilize configuration similar to those describedwithin U.S. patent application Ser. No. 14/882,085, entitled Applicationof Orbital Angular Momentum to Fiber, F30 and RF, filed Oct. 13, 2015which is incorporated herein by reference in its entirety. These variousimplementations are more fully described hereinbelow. This technique maybe implemented into the full duplex communications system describedabove.

Referring now also to FIG. 59, there is illustrated a more detaileddescription of a small cell network KPI (key performance indicator)5902. The small cell network KPI 5902 is implemented within the SDNcontroller 30902 to enable communications between the SDN controller andsmall cells within the small cell network. As mentioned previously, theOpenDaylight controller 5911 provides routing infrastructure for thesmall cell operation. The OpenDaylight controller 5911 utilizes anapplication program interface 5904 for enabling communications betweenthe controller 5911 and an orchestrator 5913. The orchestrator 5913dynamically optimizes the small cell network by minimizing power andlatency while maximizing the capacity of the network. The network KPI5902 must maintain a communication channel 5907 with the SDN controller5904 in order to be able to exchange control plane messages with thesmall cell nodes 5909. This communication channel 5907 can beestablished in the same network interface as the one used for the dataplane (in-band connectivity) or in a different interface (out-of-band).With in-band connectivity, the infrastructure costs are reduced, but iflink failure occurs, the node loses the connection with the controller5911. Out-of-band control plane connectivity requires an additional NIC(network interface controller) in the managed devices. An LTE interface5911 is used on each SDN enabled small cell backhaul node 5910 for SDNcontrol plane connectivity, in order to provide a robust channel andreduce SDN control latency while the data plane is using the multi-hopbackhaul connectivity over a multiband (mmWave, sub 6 GHz and FSO)network.

Small cell wireless links may have dynamic link outages, especially whenoperating at mmWave band. A link can temporarily go from non-line ofsight to outage (e.g. due to blockage), leading to changes in thetopology and consequently, in the available capacity. When such eventshappen, the SDN controller 5903 can perform path recalculation betweenthe small cell nodes 5909 but the process may take a significant amountof time. The network KPI 5902 as illustrated in FIG. 59 uses fastfailover (FF) group tables 5906 from the OpenFlow plug-in 5908 torapidly repair link failures locally.

The orchestrator 5913 communicates with the multidimensional optimizer5910. The Application Program Interface 5904 is used to communicate withthe orchestrator 5913 in order to perform the reconfiguration of thesmall cell network. Also, this configuration can be triggered by theorchestrator 5913 through this REST API. The new configurations arepushed to the wireless communications services (WCS) and new paths arerequested to the Path Calculator. The multidimensional optimizer 5910finds a maximum value based upon latency, capacity and 1/power usingEuler-Lagrange multipliers. The network KPI 5902 further includes apacket inspection/handling module 5912. The packet inspection/handlingmodule 5912 inspects and controls the data packets that are transmittedover the communications channels 5903 to the small cell nodes 30908. Thepacket inspection/handling module 5912 parses packets sent to the SDNcontroller 5903 (e.g. for new flows when no rules are installed at thesmall cell nodes 5909). The extracted information is sent to the pathcalculator 5914, which replies with a primary path from the source tothe destination node according to a given path calculation strategy. Theoriginal packet is then sent back to the destination node.

The path calculator 5914 is responsible for calculating alternate pathsto the small cell nodes 5909 when existing links fail. The pathcalculator 5914 computes paths between the powered on small cell nodes5909 and instructs the installation of new forwarding rules. The pathcalculator 5914 uses a network graph that contains only the activenodes. If the fast failover (FF) strategy is active, a maximum disjointpath is also calculated from each intermediate node, and the requiredforwarding rules are installed in combination with the usage of the FFgroup table 5906 feature from OpenFlow. The link/path processingcalculation module 5914 uses information from the neighbor list 5916 tomake the new path calculations. The neighborhood mapper 5916 is adatabase list of small cell nodes and their associated neighboringnodes. The neighborhood mapper 5916 infers the neighborhood andinterference graph for each node/link from the existing topology. Smallcell nodes 5909 send out periodic beacons to neighbors. The collectedinformation statistics are sent to the SDN controller 5903 and used toaugment existing data from the network links.

The OpenFlow plug-in 5908 includes an OpenFlow detection module 5918 fordetecting OpenFlow messages. The flow process module 5920 calculates themessage routing. The wireless configuration service 5922 sends wirelessspecific configuration requests to the managed small cell nodes 5909through an OpenFlow protocol extension. The Wireless Statistics Manager5928 collects wireless related statistics from the managed small cellnodes 5909 over an aggregator API 5930 through an extension of thestatistics manager component 5924 from the OpenFlow Plugin 5908. Thestatistical information is fed to the statistics module 5924 from thesmall cell nodes 5909. The requests and statistics can have differenttypes, which are specified by a bit mask field in the request body. Eachof the modules within the OpenFlow plugin 5908 communicates with thesmall cell nodes 5909 through an OpenFlow API 5926. A metrics collector5928 is responsible for obtaining network performance metrics thatcannot be directly retrieved through OpenFlow plug-in 5908. The metricsare obtained through the aggregator API 5930. The calculated data ismerged into the available statistics and can be used by every other SDNcontroller 5903 component.

Due to the increasing traffic demand, existing mobile access and privatenetworks face a capacity problem. In order to increase the capacity, itis customary to deploy many small cells which may be dynamicallycontrolled based upon traffic demand as illustrated in FIG. 60. Theprivate small cell network 6002, as described previously, consist of aplurality of individual small cell nodes 6004 that are interconnectedvia communication links 6006. Each of the small cell nodes 6004 areinterconnected with each of the small cell nodes within its vicinity viaone of the communication links. Thus, for example, as illustrated inFIG. 60, node 6004 x is interconnected with each of the surroundingnodes 6004 y (in this case eight nodes) through an associatedcommunication link 6006. Thus, node 6004 x can communicate over thesmall cell network 6002 through any of the adjacent small cell networknodes 6004 y.

As a consequence, the fabric for small cell networks 6002 needs to copewith the massive increase in user demands since the laying of fiber toeach small cell node 6004 is not economically feasible. It is possibleto have mmWave based private mobile networks due to the large chunk ofspectrum that is available both in unlicensed bands (the 60 GHz and70/80 GHz bands) as well as licensed bands of 24, 28 and 39 GHz.However, due to the specific propagation characteristics of the mmWavespectrum, communications links 6006 between small cell nodes 6004 mayface challenging network outages. Additionally, a more flexible designof the network 6002 is desired in order to cope with the diversificationof service requirements.

A private small cell network architecture based on the concept ofsoftware defined networking will be able to address these issues andprovide a mmWave or MulteFire based mobile network or private network.Referring now to FIG. 61, in order to cope with the dynamics of thenetwork, the SDN control plane calculates for each small cell node 6102a primary link 6104 and a set of backup links 6106. The set of backuplinks 6106 include at least one backup link which may be utilized if theprimary link 6104 goes out. Using OpenFlow Fast Failover groups such asthose described herein above, a fast local repair of a link 6106 can beachieved leading to a resilient mesh network architecture. The proposedarchitecture leads to a lower packet loss and consequently higherthroughput data rate and better private network reliability.

Referring now to FIG. 62, network reliability may also be improvedutilizing an architecture wherein each node 6202 uses SDN-based channelestimation to multiplex between line of sight (LOS) mmWaves, non-line ofsight (NLOS) sub-6 GHz and free space optics (FSO) transmissions. Thisis achieved using a LOS mmWave transceiver 6204 for transmitting line ofsight millimeter waves 6206, an NLOS transceiver 6208 for transmittingnon-line of sight sub-6 GHz signals 6210 and a FSO transceiver 6212 fortransmitting FSO signals 6214. Multiplexing control circuitry 6216multiplexes between the LOS mmWave transceiver 6204, an NLOS transceiver6208 and an FSO transceiver 6212 based upon the environmental and systemoperating conditions. When the atmospheric conditions are good, thenetwork relies upon the FSO transceiver 6212. When atmosphericconditions become foggy or rainy, the system adaptively switches to RFLOS transceiver 6208 or the LOS transceiver 6204 using the multiplexercontrol 6216. If the operating environment has many physical obstaclesbetween the transmitter and the receiver, the system would select theNLOS transceiver 6208.

Despite introducing new technologies at lower layers of the protocolsuch as better modulation and coding schemes or coordinating multipointtransmissions, the predicted demand is much higher than what can besupported with new physical layer only technologies in the short term. Acommon assumption to provide increased capacity at scale is to use ahigher frequency band were more spectrum is available and to reduce thecell size in order to increase spatial reuse. Network operation is oftendominated by proprietary solutions which hinder innovation. An importantchallenge to solve for small cell backhaul links is an efficient butflexible forwarding architecture which relays user data over a multi-hopwireless backhaul between a plurality of small cell nodes.

Referring now to FIG. 63, in a typical SDN-based architecture, acentralized controller 6302 installs within small cell nodes flexiblerules 6304 that determine the forwarding behavior within the data plane.However, a forwarding configuration addressing the inherent resiliencychallenges to cope with unstable backhaul links using a combined RF LOS,NLOS and FSO has not been previously addressed. A resilient forwardingconfiguration of an SDN-based small cell private wireless network 6306that focuses on SDN-based resiliency mechanisms and uses the concept ofOpenFlow fast failover groups 6308 as described hereinabove. Thecontroller 6302 calculates each link 3306 for each small cell node 3302backup links toward the Gateway. The main link and the backup link areboth placed into a fast failover group 6308. The small cell node 3302uses rapid link monitoring to locally detect if a link is in the outagestage, in which case, the OpenFlow-based fast failover locally switchesfrom a main link to a backup link.

The traditional SDN concept relies on a centralized control plane, whichexercises control on forwarding decisions in the data plane.Consequently, the control and data planes are decoupled which allows avery flexible forwarding control. However, using SDN for small celllinks present several challenges. This is because the performance andreliability of mesh-based networks such as that illustrated in FIG. 60depends on fast local reactions to topology changes where a centralizedcontrol plane is typically too slow to react. Therefore, there beenattempts to use proprietary routing and forwarding decisions based ondistributed protocols were SDN is used to steer traffic.

In an alternative approach as illustrated in FIG. 64, the forwardingdecisions inside the small cell network may be configured by the SDNcontrol plane. SDN-based resiliency using fast local restorations 6402(e.g. implemented inside the data plane 6404 of the cell small nodes6406) may be used whenever local problems such as link outagetransitions of the millimeter wave links are detected. This local repairmechanism, which can be preinstalled, avoids the need to ask thecontroller how to react in a case when a neighbor node cannot be reachedanymore and leads to a more robust data plane behavior.

Referring now to FIG. 65 there is illustrated the process forimplementing SDN-based local repair. The process uses SDN to calculate aprimary and a secondary path for private small cell network nodes.SDN-based local repair is implemented using fast failover groups 6308(FIG. 63). A primary and a secondary action are put into the same group.Consequently, the SDN controller calculates for each small cell aprimary path at step 6502 towards the gateway. Additionally, the SDNcontroller calculates a backup path towards the gateway at step 6504.Based upon the path calculations, the SDN controller installs forwardingrules for the primary path at step 6506 into the fast failover group6308 and will additionally install forwarding rules at step 6508 for thebackup path into the fast failover group.

Once data packets arrive at a small cell node at step 6510 which shouldbe forwarded to a neighboring small cell node using mmWave or MulteFirelinks, the data packets will be forwarded according to the first port inthe fast failover group table whose port state is active. This requiresan OpenFlow data path implementation which uses for each neighbor adedicated OpenFlow port. Once the primary port is detected to be down,the data packets are automatically forwarded using the next active port,i.e. towards a different neighbor where there is an active backup link.Thus, a determination is made at inquiry step 6502 if the primary pathis working. If so, control passes on to step 6514, and the packet isforwarded on the primary path. If inquiry step 6512 determines that theprimary path is not working, the packet is forwarded on the backup pathat step 6516. The process is then completed at step 6518. This processallows the small cell node to perform a local failover instead of theSDN controller performing a centralized failover. This significantlyreduces the reaction time to failures in the mesh forwarding structure.

Referring now to FIG. 66, there is illustrated the process for detectinglink state and transmitting on primary and backup links. In order todetect the link state, bidirectional forwarding detection (BFD) is acommonly used technology. BFD determines the state of the port byestablishing a connection at step 6602 using a three-way handshakeroutine. BFD next waits a predetermined period of time at step 6604 andtransmits a periodic control message at step 6606. Inquiry step 6608determines if a response to the control message has been received. Thetimeout period is determined by the control messages between the BFDmessages. If a response to the control message is received,transmissions are carried out on the link at step 6610 and controlpasses back to step 6604 to await an additional predetermined period. Ifno response to the control message is received within a specified timeinterval, the link is considered down at step 6612. In this manner, linkfailures may be rapidly detected and reacted to. Consequently, smallcell backhaul nodes would send periodic BFD messages to each neighboringnode over the connection links to detect link states. Once BFD detects alink down event at inquiry step 6608, the link state is set to down atstep 6612. This triggers the OpenFlow datapath to start transmitting toa different neighbor small cell. This is achieved by selecting a backuplink at step 6614.

Alternatively, MAC layer protocol messages can be used in order to inferthe state of the communications links, which could be integrated intothe OpenFlow data path. The media access control (MAC) layer is a lowersublayer of the data link layer of the seven-layer OSI model. The OpenSystems Interconnection model (OSI model) is a conceptual model thatcharacterizes and standardizes the communication functions of atelecommunication or computing system without regard to their underlyinginternal structure and technology. Its goal is the interoperability ofdiverse communication systems with standard protocols. The modelpartitions a communication system into abstraction layers. The originalversion of the model defined seven layers.

A layer serves the layer above it and is served by the layer below it.For example, a layer that provides error-free communications across anetwork provides the path needed by applications above it, while itcalls the next lower layer to send and receive packets that comprise thecontents of that path. Two instances at the same layer are visualized asconnected by a horizontal connection in that layer.

The MAC sublayer provides addressing in channel access controlmechanisms that make it possible for several terminals or network nodesto communicate with a multiple access network and incorporates a sharedmedium, e.g. an ethernet network. The hardware that implements the MACis referred to as the media access controller. The MAC sublayer acts asthe logical link controller (LLC) sublayer and the networks physicallayer. The MAC layer emulates a full-duplex logical communicationchannel in a multipoint network. The channel may provide unicast,multicast or broadcast communication service.

A weakness with fast failover is that it can only perform localfailover. If no alternative local path is available, e.g. all neighborsare not reachable anymore, then crankback routing must be performed.This requires that the packet be sent backwards toward the source to asmall cell node which has an alternative active path towards thedestination. Thus, crankback forwarding can potentially have largeimpacts on the latency. Such latency can be reduced significantly byintroducing stateful forwarding in the data plane using OpenState.OpenState is a research effort focused in the development of a statefuldata plane API for Software-Defined Networking. OpenState is moreparticularly described in the OpenState v1.0 specification. If packetsarrive at a small cell forwarding node which does not have a next hoptowards the destination node because the link is down, the node tags thepacket and the packet is sent back towards the source. When the messagereaches the small cell node having a backup path, the state of theforwarding rules change in such a way that the coming packet traversethe backup path already at the node. Therefore, once a backup link isselected at step 6614, inquiry step 6616 determines if the backup linkis operating and available. If so, transmissions are carried out on thebackup link at step 6618 and control passes back to step 6604. If thebackup link is not available as determined at inquiry step 6616, thecrankback forwarding process is carried out at step 6620 andtransmissions carried out on allocated available link.

Referring now to FIG. 67, the use of SDN and NFV within the edge/accessnetwork 6702 connected to a backhaul network 6704 in addition to beinguseful as described herein above, may also be used for providing a highcapacity last drop access connections to users 6706 that lowers costsand guarantees flexibility. Last drop connection comprises a wirelessaccess 6702 from the edge/access network 6704 to the user 6706. The lastdrop connection 6708 is the access connection to a network by user 6706.Data from a user 6706 may be provided over the wireless last dropconnection 6708 to a private network 6707 and then on to the edgenetwork 6702 and forwarded onward to the backhaul network 6704 andnetwork core 6710. These last drop connections 6708 can be providedbetween the private network 6707 and users 6706 in a number of fashions.A last drop connection 6708 provided in the described manner provides ahigh-capacity access solution that lowers cost and guaranteesflexibility and scalability for both residential and enterprisecustomers with 60 GHz access in both backhaul and fronthaul.

Mesh networks can be applied to both access networks 6702, 6707 (frombase to end users) and backhaul networks 6704 (from base to networkcore). A mesh network means that each node is connected to at least twoor more sites, so that when a link is broken, the mesh network canself-heal itself, by finding another path to keep the networkconnections active. FIG. 68 illustrates the use of a mesh network 6802for the last drop connections 6708 between the user 6706 and the edgenetwork 6702. While the following description is made with respect tousing a mesh network to provide the access network 6702 connections, themesh network could be applied in a similar manner as part of thebackhaul network 6704. The last drop connection 6708 comprises what maybe termed as a self-organized network for access 6902 as illustrated inFIG. 69. The mesh network 6802 may comprise one or both of an indoor oroutdoor network. The mesh network 6802 uses an SDR based indoor andoutdoor MulteFire system to allow a private network to provide scalablecoverage, capacity and control that is not available in public networks.The self-organized network for access 6902 includes a number ofcharacteristics enabling the communications. The self-organized network6902 includes a number of characteristics including that the network isself-healing 6904, self-configuring 6906, self-optimizing 6908 andprovides auto provisioning 6910. The network 6902 is self-healing 6904in that when a communications link within the network 6904 breaks down,the network may self-correct the problem in order to cure the failedlink. The network 6902 is self-configuring 6906 in that the networksoftware may reconfigure the links automatically without any externalinputs from a network manager in order to correct failed communicationlinks. The network 6902 is self-optimizing 30908 in that decisions forplacement of nodes within the network 6902 are made in order to optimizethe closeness to the various end-users and to further minimize backhaulcost in order to reduce overall network costs. Finally, the autoprovisioning functionalities 6910 of the network 6902 enable the networkto automatically establish new communication links that are generatedbased upon failed existing communication links.

Various optimization techniques may be utilized. An effective hybridtechnique may be used for optimizing multimodal functions in large scaleglobal optimization (LSGO) that will pair the first search spaceexploration performed initially by standard techniques with moreefficient local search techniques. Large scale global optimization(LSGO) is as important technique in large scale traffic networks. Asdimensionality increases, the performance of most optimizationalgorithms quickly goes down. There are two major reasons for thisdecrease in performance. These are an increase of the complexity and anexponential increase of the search space volume. Due to the increase incomplexity, unimodal functions may become multimodal in largedimensions. Due to the exponential increase of the search space volume,optimization algorithms need to increase their efficiency when exploringlarge search spaces. The efficiency can be measure by the number offunction evaluations required to converge to a given optimum. Inpractice, many large-scale problems are multimodal.

In addition to the exponential increase in the number of candidatesolutions, the cost of converging to any local optimum also increases.In high dimensional search spaces, we must focus almost exclusively ongradient exploitation in order to guarantee convergence to any localoptima. However, disregarding exploration may lead to poor results inmultimodal problems. In multimodal problems, it is critical to explorethe search space to find the most promising regions before convergingtoward a local optima. Even in LSGO some exploration is necessary toachieve good performance on multimodal problems. There is a need tofocus on minimum Population Search. The key idea behind the approach isto focus on multi-modal functions and to consider from the beginning theissues when scaling to large scale global optimization. This is done viaan efficient use of function evaluations and an unbiased exploration.

In the current approach, search techniques focus more and more ongradient exploitation as dimensionality increases. So the primary focusis on hybrid techniques which will pair the full search spaceexploration performed initially by standard techniques with moreefficient local search techniques. Therefore, an effective hybridtechnique is used for optimizing multimodal functions in LSGO.

Minimum Population Search focuses on multi-modal functions. Originallythe ideas were developed for two dimensional problems, later generalizedfor standard dimensions and scaled towards large scale problems.Standard techniques perform a methodical and unbiased exploration basedon the Thresheld Convergence (TC) technique. Threshold Convergence isdesigned to avoid a biased exploration by preventing global and localsearch steps from happening at the same time. This is achieved by fixinga minimum search step (threshold) which decays as the search progresses.Convergence is thus “held” back until the last stages of the searchprocess.

An iterative optimization procedures built around the concept ofself-adaptation called Covariance Matrix Adaptation (CMA) with (μ, A)selection considers the best μ solutions out of a population with Asolutions for recombination. It is an iterative optimization proceduresbuilt around the concept of self-adaptation. The parameters of thesearch strategy evolve together with the solutions. CMA is an evolutionstrategy with (μ, A) selection considers the best μ solutions out of apopulation with A solutions for recombination. Recombination operatorsthen create a (single) parent representation from the μ selectedsolutions, and A new children are produced through the use of aprobabilistic mutation distribution. CMA-ES (Evolutionary Strategy) usesparameterized multivariate normal distribution for the representation ofthe mutation distribution.

A hybrid method is used for the optimization of multi-modal problems byidentifying promising attraction basins and finding the local optima inthese basins. The optimization of multi-modal problems involves twotasks including the identifying promising attraction basins and findingthe local optima in these basins. To effectively perform each of thesetasks, different search strategies may be used. The hybrid technique ofstandard MPS takes care of this issue by assigning a differentheuristics to each task. MPS's ability to efficiently explore the searchspace is used during the early stages to identify promising attractionbasins.

Referring now to FIG. 70, there is illustrated a portion of a meshnetwork 7002 used for implementing the self-organized network for access6902 described with respect to FIG. 69. The mesh network 7002 includes aplurality of nodes 7004. Each of the nodes 7004 include at least twolinks 7006 providing a pathway from the node to at least two othernodes. This is done to provide a backup link should a primary link fail.This use of the backup and primary link utilizes SDN and NFV processessuch as those described hereinabove. The wireless communication links7006 between nodes are provided using standard-based unlicensed V-band(60 GHz) or MulteFire frequencies (3.5-5 GHz) to provide SDN/NFV basedpacket network communications between the nodes 7004. This provides fora 1 Gbps access to users. The use of SDN and NFV processes are used formaintaining the communication links 7006 between nodes 7004. Thedescribed system provides solutions to support existing and futuretraffic demands using a system that leverages existing technologies, newprocesses, topologies and architectures.

FIG. 71 illustrates a mesh network 7102 that may be used forinterconnecting with a number of users. The mesh network 7102 comprisesa number of nodes 7104 that communicate with each other using V-band (60GHz) or MulteFire (3.5-5 GHz) communications transceiver. The 60 GHzV-band has been standardize under the WiGig standards. The 60 GHz V-bandcurrently provides 7 GHz of spectrum. However, the US is planning toexpand the V-band spectrum to include an additional 7 GHz to provide atotal of 14 GHz of spectrum. The 60 GHz V-band spectrum provides acommunications between nodes at a distance of approximately 200 m to 250m from each other as indicated generally at 7106. Thus, the transmissiondistance between nodes 7104 within the 60 GHz system is somewhatlimited. The MulteFire frequencies are in the 3.5-5 GHz frequency range.The mesh network 7102 implements a number of phase array antennas at thenodes 7104 to retain the highly directional signal required for 60 GHzor 3.5-5 GHz, but makes the communication links steerable to communicateover a wide area. The use of the 60 GHz V-band or 3.5-5 GHz band enablestransmitted signals to be routed and steered around interferencetypically found in dense urban environments, such as tall buildings orinternet congestion due to high user traffic. Thus, as illustratedgenerally at 7108 by the dashed line, an interfering structure orphenomena, such as a building, may prevent signals from being easilytransmitted from nodes positioned at locations 7104A. Thus, whentransmitting from node 7104B to node 7104C, system controllers wouldutilize nodes 7104D rather than nodes 7104A to route signals to steeraround the interference structure 7108.

Thus, as shown in FIG. 72 a mesh network would enable an interconnectionbetween a network site 7202 and a number of residential or enterpriseusers 7204. In this case, each of the access sites 7206 comprises nodeswithin an associated mesh network. The access sites 7206 wirelesslycommunicate with each other over communication links 7208 andcommunicate with the network sites over communication links 7210 whichmay be wireline or wireless. The access sites 7206 communicate with theresidential or enterprise users 7204 over a 60 GHz or 3.5-5 GHz wirelesslink 7212.

The communications over the wireless communications link 7212 betweenthe access sites 7206 and in the users 7204 may be implemented using MAClayer protocols and TCP-IP protocols. Normally, MAC layer protocols andTCP-IP protocols are used for packet data transmissions over wirelinenetworks. However, modified, high performance MAC layer protocols(TDMA-TDD MAC) and TCP-IP protocols may be used for communicating over awireless communications link such as that utilize between the accesssites 7206 and users 7204 that overcome the shortcomings of TCP-IP overa wireless link. This can provide up to a 6× improvement in networkefficiency and make the TCP-IP protocol more predictable on a wirelesslink as compared to the existing Wi-Fi/WiGig standard. One example of amodified MAC layer protocol and TCP-IP protocol that may be utilized forwireless communications has been implemented by Facebook. Facebook isimplemented using IPv6-only nodes and an SDN-like cloud computecontroller and a new modular routing protocol for fast root convergenceand failure detection. The Facebook system has re-architected the MAClayer protocol to solve the shortcomings of TCP-IP over a wirelesscommunications link. The modified MAC layer protocol called TDMA-TDD MACimproves wireless network efficiency by 6X. By using concepts derivedfrom LTE-TDD in a 60 GHz WiGi protocol network efficiency may beimproved. These same MAC layer implementations may be used forcontrolling communication in the mesh network wireless communications.

The Facebook system implements a base station having 96 antennas thatcan support up to 24 different data streams simultaneously over theavailable bandwidth. The system has demonstrated a 71 bps/Hz data ratewhich will soon be increased to 100+bps/Hz. The system comprises amassive MIMO system providing spatial multiplexing that achieves 1.05Gbps bidirectional data throughput (2.1 Gb per second total throughputor distribution node) in the point to point transmission mode for nodesup to 250 m away. This enables up to 8.4 Gbps of total traffic perinstallation point assuming four sectors.

Referring now to FIG. 73, there is illustrated a configuration of theaccess and network sites that are used for providing the self-organizednetwork for providing access to a number of user locations. The networksite 7302 comprises an antenna 7304 communicating with the access site7306 over a point-to-point E-Band link 7308. The antenna array 7304 mayin one embodiment utilize Fujitsu GX 4000 antennas. The point-to-pointE-band link 7308 comprises a 70 GHz backhaul link for interconnectingthe network site 7302 with the access site 7306. The network site 7302also includes a mini data center 7310 for storing data that may beaccessed by users 7312. Data may be uploaded to or downloaded from themany data centers 7310 over an associated fiber ring 7314interconnecting the network site 7302 to other network sites.

The access site 7306 also includes an antenna 7316 for providing thepoint-to-multipoint E-band and link 7308 with the network site 7302. Theaccess site antenna 7316 may also comprise in one embodiment a FujitsuGX 4000 antenna. The access site 7306 additionally includes a phasedarray of antennas 7318 for providing a point-to-multipoint V-bandconnection 7320. The phased array antennas 7316 providepoint-to-multipoint transmissions to a plurality of residential orenterprise users 7312. Placement of each of the access site 7306 andnetwork site 7302 are achieved using the Optima System that optimallylocates access and network sites that is closest to the end-user butminimizes backhaul cost. At the access sites, specially constructedmini-towers provide a high-capacity, last drop access solution thatlowers cost and guarantees flexibility. Placement of the access sites7306 in this manner using the Optima System maximizes operation of thenetwork. The Optima system determines the optimal longitude and latitudefor the access site and additionally includes the Z value (height) ofthe antennas 7318 and 7316 in order to best locate the antennas on anaccess site poll. While the illustration of FIG. 73 illustrates a singleaccess site 7306, it will be appreciated that the mesh network systemwill comprise a plurality of access sites 7306 each comprising aspecially structured mini-tower.

The access sites 7306 and network sites 7302 may be configured to enableenhanced fixed broadband (eFBB), ultra-reliable low latencycommunications (uRLLC) with massive MIMO transmissions. The massive MIMOtransmissions are provided from antenna arrays at access sites 7306 andnetwork sites 7302 that provide for multiple input/multiple outputtransmissions. The antenna arrays from the access sites 7306 and thenetwork sites 7302 provide for multipoint-to-multipoint andpoint-to-point transmissions. Enhanced fixed broadband is a fixedbroadband that is enhanced with new advances (i.e., New Radio (NR)technology that uses new 3GPP advances in spectral mask that are morelocalized) as well as higher order modulation and new spectrum withcarrier aggregation. Ultra-reliable low latency communications comprisesa service category designed to meet delay-sensitive services such as thetactile Internet, vehicular to vehicular communication, autonomousdriving and remote control. uRLLC has a time-to-transmit latency time(the time required to transmit a packet) of not greater than 0.5 ms. Theperformance level of a uRLLC system should provide a block error rate ofat least 10⁻⁵.

Referring now to FIG. 74, there is provided a broader level networkillustration of a plurality of optimally located access sites 7306 andnetwork sites 7302 using both licensed and unlicensed data bandscomprising a private MulteFire network. In this case, the network sites7302 comprise mini-towers with associated antennas located on MDUs(multiple dwelling units). Each of the mini-towers 7402 can eithercommunicate via both the 28 GHz licensed band point-to-point link 7404to access sites 7306, to other network sites 7302 via point-to-pointe-band connections 7406 or 3.5-5 GHz band. The network site 7302communicates with the access sites 7306 via the 28 GHz licensed bandpoint-to-point links 7404. The network sites 7302 can communicate withother network sites 7302 by either a fiber link 7408 or with othernetwork sites 7302 through a 70-80 GHz e-band point-to-point link 7406.In addition to communicating with other network sites 7302 throughe-band point-to-point links 7406, a network site may utilize the e-Bandand point to point links 7406 to communicate with connected sites 7410and smart city sites 7412.

Smart city sites 7412 comprise data collection sensors for supplyinginformation used to manage assets and resources efficiently within anurban area. The data collected is processed and analyzed to perform avariety of functions including monitoring and managing traffic andtransportation systems, power plants, water supply networks, wastemanagement, law enforcement, information systems, schools, hospitals andother community services. The connected sites 7410 comprise othernetwork data transmission or collection sites that may be utilized bynetwork site 7404. The network sites 7302 may also connect via apoint-to-point V-band connection link 7414 to X-sites 7416. An X-sitecomprises other sites that may have surveillance cameras or sensors fordifferent applications such as detection of gas for emergencies, etc.Finally, the network sites 7202 may connect to other MDUs 7418 using apoint-to-multipoint V-band access link 7420.

Access site 7206A comprises a BB (broadband) access site from amini-tower 7422 located on an MDU 7424. A BB access sites are forconnecting the end user to the access point (base station). These sitesare differentiated from aggregation sites and their associated backhaul(from an aggregation point towards the core of the network rather thantowards the end users). The access site 7206 provides point tomultipoint 60 GHz V-band or 3.5-5 GHz band links 7426 to multiple userlocations 7428. Access site 7206B comprises a mini-tower 7430 providingpoint-to-multipoint 60 GHz V-band or 3.5-5 GHzband links 7426 tomultiple users 7428.

Referring now to FIG. 75, the private system 7502 described hereinbuilds upon existing technologies in a unique combination along with newtechniques in order to provide the unique last drop system. The WiGigprotocol 7504 implemented within the Peraso chipset provides forwireless packet data transmissions. The modified MAC/TCP-IP protocol7506 that has been developed by Facebook is utilized for providingimproved wireless packet data transmissions. Communications between theaccess sites and network sites of the mesh network are controlledutilizing mesh software 7508. The mesh software 7508 is responsible forautomatically detecting when a link goes down between nodes within themesh network and reconfiguring and reestablishing a connection using anew link path. Connections between access nodes of the mesh network andusers are provided using point to multipoint transmission techniques7510 from the access nodes to the users. The control of the mesh networknode connections are carried out using SDN/NFV software controltechniques 7512 as described herein.

FIG. 76 illustrates the functional blocks of a mesh network used forproviding the last drop services to users. The mesh network structureutilizes a central controller 7602 and a mesh client 7604 that operateon top of a WiGig baseband layer 7606. The central controller 7602provides for the establishment of links between mesh network nodes andcontrols the reestablishment of failed links when necessary. The meshclient 7604 is located at each mesh network node and provides theinteractions with the central controller 7606 in order to control linkestablishment at the nodes. Each of the central controller 7602 and meshclients 7604 within the network nodes utilize the WiGig basebandcommunication control protocol 7606 in order to carry instructionsbetween the central controller 7602 and mesh clients 7604. The combineduse of the central controller 7602 and mesh clients 7604 with the WiGigprotocol 7606 enables the use of a multi-hop topology for providinglinks that travel from one point to another through multiple meshnetwork nodes. The combination additionally provides for quality ofsignal support for the links between mesh nodes and failover managementfor failed mesh node links to provide a high reliability system. Theplatform for initiating these control layers include a phased antennaarray associated with the mesh network nodes, WiGig SoC(system-on-a-chip) located on each mesh network node. Mesh softwarecontrols the node interactions using the phased antenna array and WiGigSoC.

The mesh software as illustrated in FIG. 77, includes a number offunctionalities within the mesh controller 7702 and the mesh node 7704.The system uses big data analytics for targeted decision-making, networkawareness, advanced processing with multi-core radios, dynamic porting,open source standards and network slicing. The mesh controller 7702software and mesh node 7704 software may be updated via the Internet ora private network 7706. The mesh controller 7702 provides centralizedrouting functionalities 7708. The centralized routing functionalitiesenable for centralized control of the routing of data packets overcommunication links that have been established within the system. Theneighbor selection functionalities 7710 provide for the selection ofneighboring nodes that will be used for reestablishing failedcommunication links when a primary communication link fails. The networkmetrics and status functionalities 7712 tracks network status parametersand maintains these to assist in rerouting decisions of other meshcontroller 7702 functionalities. The quality of service (QOS)functionalities 7714 monitor EIR (excess information rate) and CIR(committed information rate) traffic in order to assist in maintainingquality of service of signals being transmitted over the wirelessconnection between nodes of the mesh network for high reliability. TheOpenDayLight functionalities 7716 provide for network management anddata switching within the mesh network. OpenDaylight is a collaborativeopen source project hosted by the Linux foundation. OpenDaylightpromotes software defined networking (SDN) and network functionvirtualization (NF V).

The mesh node 7704 functionalities provide for maintenance of thenetwork of mesh nodes and providing for communications therebetween. Thebuffer management functionalities 7718 provide for buffering of datathat is being transmitted between nodes of the mesh network. The openvirtual switch (OVS) functionalities 7720 provide for switching betweennodes of the mesh network. OVS is an open source implementation of adistributed virtual multilayer switch. OVS provides a switching stackfor hardware virtualization environments while some hoarding multipleprotocols and standards used within computer networks. The ethernetconvergence layer/layer 2 bridge 7722 provides for the aggregation ofmultiple networks into a single network. Ethernet functionalities 7724provide for ethernet communications between network components over themesh network nodes. The mesh controller interface 7726 uses OpenFlow toenable communication between the various nodes of the mesh network. Thenode manager and state machine 7728 monitors the nodes of the meshnetwork and manages and tracks their state for data packet transmission.The configuration manager 7730 is responsible for configuring the meshnode network when breakdowns in nodes occur requiring reconfigurationsof links between nodes. The routing manager 7732 is responsible forgenerating routing information for packet data that is transmitted overactive links within the mesh network. The link metric functionalities7734 monitor and track the status of links between nodes of the meshnetwork. The scheduling and quality of service manager 7736 monitors thescheduling of packets between nodes and manages quality of service oflinks between nodes. The event manager 7738 is part of the mesh nodethat collects and manages all events in mesh control, node and statemachine, configuration management, routing management, scheduling andQoS management, link status and metrics, etc. The baseband hostinterface 7740 provides for an interface with the application layer bythe nodes of the mesh network.

The gateway and/or access point 7742 provides the access point addresscontrols to the ethernet. A number of WiGig BB and RF sectors 7744provide for interactions with the WiGig and RF network functionalitiesand provides for a host interface 7746, 802.11 ad physical interface7748 and 60 GHz phased array interface 7750.

Using these software functions within the mesh controller 7702 and meshnodes 7704 the mesh software implemented within processor/server at thevarious nodes can perform a number of operations within the meshnetwork. The software enables the performance of network discovery andautonomous neighbor selection. This enables nodes to identify the meshnetwork and automatically determine neighboring nodes that are locatedin close proximity to the node. The system may also perform topologymanagement using in-band signaling with the mesh network controller7702. The functions allow for the configuration of the nodeprocessor/server for management of node and sector state machines withinthe mesh network. The mesh software may also be used for detecting linkfailures and switching the communication links to an alternative pathwhen an existing link has failed. The mesh software may also be used forsector and node recovery when particular sectors or nodes within themesh network are lost. The mesh software performs link metric collectionthrough for example the link metrics functionalities 7734 to enable thedecisions and handling of things such as link failure to beappropriately decision based. The virtual switches 7720 implement L2transport. The mesh software can also provide simple network managementprotocol (SNMP) using MIBs (management information bases). SNMP is anInternet standard protocol for collecting and organizing informationabout managed IP devices on IP networks in for modifying thatinformation to change device behavior. SNMP is widely used in networkmanagement for network monitoring. SNMP exposes management data in theforms of variables on the manage systems organized in a managementinformation base which describe the system status and configuration.These variables can then be remotely queried by managing applications.Finally, the mesh software provides for ease of configurability via theuse of text file configuration parameters.

As discussed above, baseband RF transceiver and antenna arrays have beendeveloped that generate plane waves at 60 GHz and are miniaturized on aUSB dongle. These systems have been further improved using two layers ofOAM energization on a same USB dongle to provide full duplex isolation.Referring now to FIG. 78, there is illustrated the manner for systemoperation in the full duplex cancellation domain. Transmit bit 7802 areprovided to modulation and coding circuitry 7804 within amodulation/demodulation circuit 7806 to provide for modulation andcoding thereof. The baseband signal from the modulation and codingcircuitry 7804 is provided to a digital to analog controller converter7808. The analog baseband signal from the digital to analog converter7808 is provided to a mixing circuit 7810 to be mixed with a carriersignal provided by oscillator 7812. The mixing circuit 7810 provides anRF band signal to the high-pass amplifier 7814. The RF signal istransmitted from a multilevel patch antenna array 7816 as a transmittedsignal 7818.

In addition to generating transmitted signals 7818, the multilevel patchantenna array 7816 also receives signals 7820. The receive signals 7820are provided to a summation circuit 7822 that are combined with acancellation signal from a cancellation circuit 7822. The cancellationcircuit 7824 generates a cancellation signal responsive to the output ofthe high-pass filter 7814 that is being provided to the multilevel patchantenna array 7816 and a cancellation control signal 7826 provided fromthe modulator/demodulator 7806. The cancel signal from the summationcircuit 7022 is provided to the input of a low noise amplifier 7830 thatis down converted from the RF domain to the baseband domain using amixing circuit 7832 that combines the output of the low noise amplifier7830 within carrier signal from oscillator 7814. The output of themixing circuit 7832 is provided to a analog-to-digital converter 7834this digital signal is provided to the demodulation and decodingcircuitry 7836 within the modulator/demodulator 7806. The demodulationand decoding circuitry 7836 outputs the received bits 7838. Signalswithin the modulator/demodulator 7806 up to the digital to analogconverter 7808 and analog-to-digital converter 7834 comprise the digitaldomain 7840. The signals between the digital to analog converter 7808and the analog-to-digital converter 7834 and the multilevel patchantenna array 7816 are within the analog domain 7842. Finally, thesignals from the multilevel patch antenna array 78 and 16 outward are inthe propagation domain 7844.

Referring now to FIG. 79, there is more particularly illustrated fullduplex cancellation techniques for canceling interference in full duplextransmissions. Input signals x₁(1) and x₁(s) are provided to and inverseFast Fourier Transform (IFFT) 7902. The parallel outputs of the IFFT7902 are provided to a parallel to serial converter (P/S) 7904. The P/S7904 provides a serial output to an add prefix circuit 7906 thatprovides a prefix to the serial signal. The signal is provided to adigital to analog converter 7908 to convert the digital signal to ananalog signal. The generated analog signal is transmitted by atransmitting radio 7910 using a multilevel patch antenna array 7912.

Signals received by the multilevel patch antenna array 7912 are providedto a summation circuit 7914 along with a cancellation signal 7916. Thecancellation signal 7916 is generated from a cancellation control signal7918 that is provided to an inverse Fast Fourier Transform (IFFT) 7920.The output of the IFFT 7920 is provided to a parallel to serialconverter 7922 and the output of the parallel to serial converter isprovided to an add prefix circuit 7924. The add prefix circuit 7924 addsa prefix to the serial signal and provides the signal to a digital toanalog converter 7926. The digital signal passes through a transmitradio 7928 to provide the cancellation signal 7916.

The signals from the summation circuit 7914 are provided to a receiveradio 7930. The output of the receive radio 7930 is provided to ananalog-to-digital converter 7932. The digital output is provided to aremove prefix circuit 7934 to remove the prefix from the signal andforwarded to a serial to parallel converter 7936. The serial to parallelconverter 7936 converts the serial signal from the remove prefixcircuitry 7934 into parallel signals that are provided to the FastFourier Transform (FFT) 7938. The FFT 7938 generates the output signalsy₁(1) and x₁(s) from the digital signals.

Referring now to FIG. 80, there is illustrated the manner in which thepatch antennas 8002 are vertically separated from each other to limitinterference therebetween. The vertical separation can be seen both inthe transmissions between a first patch antenna array 8004 and a secondpatch antenna array 8006.

There are number of issues with respect to full duplex transmissions inmmWaves. Propagation, absorption (oxygen) and penetration issues are afirst group of problems associated with mmWaves. Some of these channelimpairments can be combatted using beam-forming gains. Beam-formingrequires antenna arrays with large number patch elements. However,mmWaves have short wavelengths, and therefore have no need for huge realestate. Thus, massive MIMO is needed for mmWave networks. However, thelarge number of signal with each antenna element increases the cost andpower consumption of the antenna. Therefore, it is not a good idea tobuild a fully-digital beamforming system where an individual RFfrequency chain is dedicated to each antenna. There is a need to reducethe overall hardware complexity by reducing the total number of RFchains in the mmWave transmission system. A hybrid analog-digital systemis the best where an individual RF chain is associated with eachsub-array of antennas rather than a single antenna of the array. Awideband precoding using principal component analysis can be used forthe system.

As will be more fully described herein below, one manner for overcomingthese mmWave issues is through the use of relays/repeaters. MillimeterWaves are sensitive to NLOS situations due to penetration loss caused byblockages. A relay-based mmWave system can fix the propagation rangeissue as well as quality, and reliability of transmissions. Relays havebeen proposed for variety of networks including cellular,device-to-device, and indoor applications. There are two types of relaysregenerative relays and non-regenerative relays. Conventional relayswork in half-duplex (HD) using separate frequencies or time slots. Thereare 2-way HD amplify-forward (AF) relay-based systems where thetransceiver and relay filters are found by solving asum-mean-square-error (SMSE) optimization problem for mmWave channel.One solution for overcoming the mmWave issues would be to have acombined precoder and relay filter matrices in HD for mmWavetransmissions. An alternative solution would be to have a hybridbeamforming for HD relay-based mmWave system with multiple receivers.

With a massive MIMO Full Duplex (FD) relay system, small distancebetween antenna patches in mmWave system creates higherself-interference within the patch antenna array. Though there have beenimprovements in FD cancelation techniques, the self-interferencecancellation for mmWave does not completely cancel the self-interferenceand there are left some residue self-interference issues within thesignal. Thus, as shown in FIG. 81, the self-interferance cancellationtechniques 8102 can be combined with the OAM orthogonality techniques8104 as described above to further suppress the residualself-interference.

Using these techniques, a hybrid digital-analog transceiver and relayfilter architecture for an in-band bi-directional relay-based FD with AFfor mmWaves can be built. The MIMO techniques described in U.S. Pat. No.10,757,576, entitled SDR-BASED MASSIVE MIMO WITH V-RAN CLOUDARCHITECTURE AND SDN-BASED NETWORK SLICING, which is incorporated hereinby reference can use imperfect channel state information (CSI) due toestimation error, feedback delays, quantization errors, pilotcontamination and perform SVD. This will provide a robust system that isresilient to imperfect channel conditions. The impact of the CSI erroris lessened by a proper precoder, receiver, and relay filter matrix

Further improvements in relay system in cancellation may be achieved maybe achieved using AI-based digital cancellation. With a massive MIMOFull Duplex (FD) relay, small distances between antenna patches inmmWave system creates higher self-interference. Analog cancellation isgenerally expensive due to the additional analog circuitry and aresidual Self-Interference (SI) signal typically still remains at thereceiver, which is then canceled in the digital domain. This requiresmodeling the non-linear effects of the different stages of thetransceiver, such as digital-to-analog converter (DAC) and ADCnon-linearities, IQ imbalance, phase-noise, and power amplifier (PA)non-linearities. Traditionally, this has been done using polynomialmodels, which have been shown to work well in practice. However, thepolynomial models have a high implementation complexity as the number ofestimated parameters grows rapidly with the maximum considerednonlinearity order. As an alternative to polynomial models, neuralnetworks (NNs) or AI-based techniques can be used for SI cancellationwhere NNs can achieve similar SI cancellation performance with apolynomial model with significantly lower computational complexity. TheAI techniques described herein above may be used for this process.

For an OFDM system, the self-interference signal received by receivepath antennas on subcarrier(s) is:I=total #of Tx patches Tx antenna i patchesJ=total #of Rx patches Rx antenna j patches

${{y_{j}(s)} = {\sum\limits_{i = 1}^{I}{{h_{ij}(s)}{x_{i}(s)}}}}{{z_{j}(s)} = {{- {h_{j}(s)}}{\sum\limits_{i = 1}^{I}{{a_{ij}(s)}{x_{i}(s)}}}}}$

After analog cancellation, the self-interference at antenna j is:

${y_{j}^{AC}(s)} = {{{y_{j}(s)} - {z_{j}(s)}} = {\sum\limits_{i = 1}^{I}{\left\lbrack {{h_{ij}(s)} - {{h_{j}(s)}{a_{ij}(s)}}} \right\rbrack{x_{i}(s)}}}}$

Perfect cancellation occurs when:

${a_{ij}(s)} = {\frac{h_{ij}(s)}{h_{j}(s)} \approx \frac{{\overset{\hat{}}{h}}_{ij}(s)}{{\overset{\hat{}}{h}}_{j}(s)}}$

Referring now to FIGS. 82 and 83 there are illustrated manners forimproving isolation using orthogonal Laguerre Gaussian (LG) modes.Previous array configurations 8202 provide a circular configuration ofpatch antennas in two concentric circles. The inner circular arraycomprises patch antennas 8204 that are used for transmitting signals.The outer circular array comprises patch antennas 8206 that are used forreceiving signals. The inner transmission array has a radius of 32 mmand the outer reception array has a radius of 60 mm. FIGS. 82 and 83illustrate different configurations with respect to rotation,orientation and offset of the arrays. In each case, the difference inthe received voltage is determined due to modes of the arrays.

With respect to FIG. 82, there is illustrated the rotation of thereceiver circular array at 8208. Within the array 8208, the outercircular receiver array comprised of patch antennas 8206 are rotated 45°about the Z axis (the axis coming directly out perpendicular to thefigure) with respect to the inner circular patch antenna arrayconsisting of patch antennas 8204. While FIG. 82 illustrates rotation ofthe outer receive array by 45°, other rotation angles such as 0°, 90°,etc. may also be utilized.

Referring now to FIG. 83, the orientation of the receiver arraycomprised of the circular array of patch antennas 8206 may also bechanged in order to provide improved signal isolation between thetransmit and receiver arrays. In the example illustrated generally at8302, as compared with 8202, the receiver patch antennas 8206 arearranged differently while maintaining the radius and spacing the sameas in the patch antenna array 8202. The array is then rotated as before.Thus, rather than having a pair of patch antennas 8206 at the top of thearray as illustrated at 8202 only a single patch antenna 8206 is locatedat the top of the orientation illustrated at 8302.

Referring now to FIG. 84, there is illustrated a table of variouseffects of rotating the circular receiver array as shown in FIG. 82 atvarious angles from 0° to 90° for two different orientations of theouter receiver circular array of patch antennas. As can be seen, thereceived voltage for OAM values of +1 and −1 changes to varying degreesbased upon the rotation.

Referring now to FIG. 85, there is illustrated a further embodimentwherein the circular receiver array utilizes 12 patch antennas 8502rather than the 8 patch antennas previously illustrated while keepingthe same radius for the circular receive antenna array. Illustration8506 illustrates the antenna with the circular receiver array rotated0°, and illustration 8508 illustrates the circular antenna array rotatedat 45°. It will be appreciated that rotation at any number of degreesand location of the antennas in various orientations in a similar mannerto that previously discussed with respect to FIGS. 82 and 83 may beutilized. Referring now to FIG. 86, there is illustrated a table ofvarious effects of rotating the circular receiver array including 12patch antennas as shown in FIG. 85 at various angles from 0° to 90° fortwo different orientations of the outer receiver circular array of 12patch antennas. As can be seen, the received voltage changes for OAMvalues of +1 and −1 to varying degrees based upon the rotation.

Referring now to FIG. 87, there is illustrated a further embodimentwherein the circular receiver array is placed on a separate substrate8704 that is vertically separated from the circular transmission array8706 located on a second substrate 8708. Each of the transmittersubstrate 8708 and the receiver substrate 8704 include a number of patchantennas thereon. The transmitter substrate 8708 and the receiversubstrate 8704 are separated at distances varying from 0 mm to 50 mm.FIG. 88 illustrates a chart of the received voltage for OAM values of +1and −1 with respect to the circular receiver array, and the circulartransmission array being separated by 0 mm (concentric) and atseparation distances of 10, 20, 30, 40 and 50 mm. The table alsoillustrates the received voltage for OAM values of +1 and −1 when thetransmit and receive arrays are separated by 10 mm and the receiverarray has 90° of rotation.

FIGS. 89-106 illustrate voltage plots for various configurations of thecircular transmitter array and the circular receiver array havingdifferent rotations, orientations, vertical separations and numbers ofreceiver array patch antennas. Each figure provides an illustration ofthe particular implementation of the combined circular transmitter arrayand circular receiver array along with a first voltage plot illustratingthe voltage for the same OAM modes, and a second voltage plot indicatingthe voltage for opposite OAM modes. FIGS. 89-100 illustrate variousorientations rotations and numbers of antennas wherein the circulartransmitter array in the circular receiver array are concentric with thecircular transmitter array located within the circular receiver array.FIGS. 101-105 illustrate voltage plots for configurations of circulartransmitter arrays and circular receiver arrays separated by distancesof 10 mm (FIG. 101), 30 mm (FIG. 102), 40 mm (FIG. 103), 50 mm (FIG.104), 20 mm (FIG. 105). Finally, FIG. 106 illustrates plots with respectto the circular transmitter array and circular receiver array beingseparated by 10 mm and the circular receiver array having a 90°rotation.

A link budget involves an accounting of all of the power gains andlosses that a communication signal experiences in a telecommunicationlink from point to point as passing from a transmitter, through a medium(free space, cable, waveguide, fiber, etc.) to the receiver. The linkbudget is defined by an equation giving the received power from thetransmitter power, after the attenuation of the transmitted signal dueto propagation, as well as the antenna gains and feedline and otherlosses, and amplification of the signal in the receiver or any repeatersit passes through. A link budget is a design aid, calculated during thedesign of a communication system to determine the received power, toensure that the information is received intelligibly with an adequatesignal-to-noise ratio. Randomly varying channel gains such as fading aretaken into account by adding some margin depending on the anticipatedseverity of its effects. The amount of margin required can be reduced bythe use of mitigating techniques such as antenna diversity or frequencyhopping. A simple link budget equation looks involves:Received power (dB)=transmitted power (dB)+gains (dB)−losses (dB)

Referring now to FIG. 107 there is illustrated a link budget for atransmitting and receiving system. The transmission power P_(tx) ismultiplied by the transmission gain Gt to provide the total gain of thetransmitted signal of 85 dBm. The signal will experience a gain loss Lof 85 dBm in transmission. The received signal has a gain boost (G_(r))applied at the receiver to provide the received power (P_(r)). Thereceived power P_(r) must be above the receiver sensitivity level of−120 dB.At 1 km the Pt+L=−85 dBm (from graph)Gr=Gain of dish=30 dB

The receiver sensitivity must be greater than the total power of thetransmission from the receiver minus the transmission losses and thegain of the receiver as shown by:P _(tx) G−L+Gr>Rx Sensitivity−85+30=−55 dBm>−120 dBmThus, the receiver must have a sensitivity of at least −120 dBm todetect the received signals. Examples of various transmission powers areshown in FIG. 108.

The link budget equation can be further defined as follows. Transmissionequation for non-zero mode orders

${\frac{P_{r}}{P_{t}}(/)} = {{G_{t_{eq}}(/)}{G_{r_{eq}}(/)}{\frac{1}{L_{FS_{eq}}(/)}.}}$where G_(teq) and G_(req) are the equivalent antenna gains for thereceiver and transmitter, and L_(Fseq) is the equivalent free spaceloss.With values substituted

${\frac{P_{r}}{P_{t}}(/)} = {\frac{Ng_{t}}{{/}!}\left( \frac{4{\pi\left( {\pi R_{t}^{2}} \right)}}{\lambda^{2}} \right)^{/}\frac{Ng_{r}}{{/}!}\left( \frac{4{\pi\left( {\pi R_{r}^{2}} \right)}}{\lambda^{2}} \right)^{/}\left( \frac{\lambda}{4\pi D} \right)^{{2{/}} + 2}}$where N is the number of array elementsg_(t), g_(r)=gain of antennasR_(t), R_(r) are the array radiil is the mode orderλ is the wavelengthD is the distance between the antennas

FIG. 109 illustrates a link budget plot in accordance with theseequations. The antenna gain P_(t)+Tx at 1 m from the antenna isindicated at 10902. The power received after the antenna gain at 0.5 kmis indicated at 10904. The power received after the antenna gain at 1 kmis indicated at 10906. The receiver sensitivity is indicated generallyat 10908.

A further necessary consideration of the repeater system is thecrosstalk efficiency. Crosstalk is a phenomenon by which a signaltransmitted on one channel of a transmission system creates an undesiredeffect in another channel or in the current system between transmit andreceive antennas. Crosstalk is usually caused by undesired capacitive,inductive, or conductive coupling from one channel to another. Crosstalkefficiency relates to the amount of crosstalk occurring betweenchannels. Referring now to FIG. 110, there is illustrated two patchantenna arrays 11002 and 11004 consisting of receiver portions Rx andtransmitter portions Tx. The arrays 11002 and 11004 are 1 km apart.Transmissions 11006 occur from a transmitter TxW to a receiver RxE andhave a power of P_(rx)E. Crosstalk may occur between the transmissionantenna TxE and the reception antenna RxE or between any othertransmitting and receiving patch antenna.

As shown in FIG. 88 with respect to the results of HPSS, the crosstalkefficiency is equal to the ΔRxV(dB) of the 10 mm separation with 90° ofrotation of the receiver array. This provides a crosstalk efficiency of29.1 dB. Crosstalk power is determined according to:P _(crosstalk)=(P _(t))ηcrossη_(dsp)=(Pt)10⁻³10⁻⁹Here η_(cross) is the efficiency of crosstalkThis is an efficiency of (29.1 dB) as obtained from HFSS simulation.η_(dsp) is the coding improvement of signal processingPt is the power transmitted=55 dBm (Tx antenna gain (30 dB) notrelevant)P _(crosstalk)=(P _(t))η_(cross)η_(dsp)=55−29.1−90=−64.1 dB

FIG. 111 illustrates a link budget for transmitted signals and crosstalkissues according to conservative estimates. Pt 11102 indicates thesignal transmission power. Pt+Tx 11104 indicates the antenna gain at 1m. There is 29.1 dB of physical crosstalk isolation 11106 between thetransmitter and receiver. There is 90 dB of crosstalk coding isolation11108. The crosstalk after adding in the gain equals −64.1 dB 11110. Thecrosstalk remains constant and is independent of distance between thetransmitter and receiver. The power at the receiving antenna afterantenna gain at 0.5 km is −45.360 dBm 11112. The power at the receivingantenna after gain at 1 km is −57.5 dBm 11114. The receiver sensitivityremains at approximately −120 dBm.

The current margins between the crosstalk power and the received powerare illustrated generally in the table below. The table illustratesvarious sensitivities at 0.5 km and 1 km. For each of these distancesthe launch power (dBm), Rx sensitivity (dBm), crosstalk power (dBm),power received (dBm) and margin (dB) are disclosed.

Distance from Tx (km) 0.5 1 Launch power (dBm) 85 85 Rx sensitivity(dBm) −120 −120 Crosstalk power (dBm) −64.1 −64.1 Power Received (dBm)−45.36 −57.5 Margin (dB) 18.74 6.6

A referring now to FIG. 112, there is illustrated a block diagram of themanner for transmitting millimeter wave signals between mmWave basestations 11202 using a repeater 11204 implemented according to theembodiments described hereinabove. A first mmWave base station 11202 atransmit signals between the base station and a repeater 11204. Thetransmitted signals may then be forwarded from the repeater 11204 to asecond mmWave station base station 11202 b. The millimeterwave repeater11204 is compatible with existing millimeterwave radios utilized by thebase stations 11202. They repeater 11202 utilizes the same frequencybands, the same normal channel plan and the same modulations from low tohigh QAM (quadrature amplitude modulation).

Referring also to FIG. 113, the repeater 11204 supports different modesand functions available in existing millimeterwave radio technologies.These include automatic transmit power control (ATPC) 11302, adaptivecoding and modulation (ACM) 11304, dual polarization (XPIC) 11306, LTEtraffic modes 11308 (data and VOLTE), video modes 11310 (digital, SD,HD, UHD, 4K), low latency 11312, frequency plans 11314 and trafficloading 11316. Automatic Transmit Power Control (ATPC) automaticallyincreases the transmit power during “fade” conditions such as heavyrainfall or other conditions diminishing signal transmissioncharacteristics. ATPC can be used separately from ACM or together tomaximize link uptime, stability and availability. When the “fade”conditions are over, the ATPC system reduces the transmit power again.This reduces the stress on the system power amplifiers and reducesoverall power consumption, heat generation and increases expectedlifetime. ACM involves the matching of modulation, coding and othersignal and protocol parameters to the conditions on the radio link.These conditions may include things such as path loss, interference dueto signals coming from other transmitters, the sensitivity of thereceiver, the available transmitter power margin, etc. XPIC, orcross-polarization interference cancelling technology, is an algorithmto suppress mutual interference between two received streams in apolarization-division multiplexing communication system.

The millimeterwave repeater 11204 may be implemented in a variety ofdifferent configurations as shown in FIGS. 114-119. FIG. 114 illustratesa single relay system wherein an enodeB 11402 transmits signals betweena repeater antenna 11404 that forwards signals on to a CPE (customerprovided equipment) receiver 11406 at a home or business location 11408.FIG. 115 illustrates a tandem configuration where in signals are relayedfrom an enodeB 11502 to customer provided equipment 11504 at a home orbusiness 11506 through a pair of repeaters 11508 and 11510. The enodeB11502 transmit signals to a first repeater 11508 which forwards thesignal to a second repeater 11510 that first repeater is operating intandem with on the communications link. The second repeater 11510transmits signals to the CPE receiver 11504 at the customer home orbusiness 11506.

FIG. 116 illustrates a Y junction configuration. The aggregated enodeB11602 transmits signals to a repeater 11604 which transmits the signalsonward further in multiple directions. The first signals are transmittedfrom the repeater 11604 to CPE receiver 11606 at a user home or business11608. A second signal is transmitted to a second CPE receiver 11610 ata second user home or business 11612. While FIG. 116 illustratestransmitting signals in two directions from the aggregated enodeB 11602to the repeater 11604 signals may be transmitted in multiple directionsas long as the multiple directionality distinction between the multipletransmitted signals may be maintained. One example of this isillustrated in FIG. 117 which illustrates an H junction configuration.In this embodiment, the aggregated enodeB 11702 transmits a signal to arepeater 11704 which transmits signals outward therefrom in threedifferent directions. Signals are transmitted to a CPE receiver at afirst home or business 11706, a second home or business 11708 and athird home or business 11710. FIG. 118 illustrates a tandem Y junctionconfiguration. The enodeB 11802 transmit signals to a first repeater11804. Repeater 11804 transmit the signal in a first direction to asecond repeater 116 and to a CPE receiver 11808 associate with acustomer home or business 11810. The signal received at the secondrepeater 11806 is further transmitted in two directions to the first andsecond CPE receivers 11808 associated with first and second customerhome or business 11812 and 11814, respectively.

FIG. 119 illustrates the configuration using tandem repeaters and apassive reflector. The enodeB 11902 transmit signals to the firstrepeater 11904 for the signal to a second transmitter 11906. The secondrepeater 11906 transmit the signal to a passive reflector 11908 whichreflects the signal toward a CPE receiver 11910 at a customer home orbusiness 11912. Applicant notes that the above comments with respect totransmitting to or from a particular enodeB, repeater or CPE receiverinvolves bidirectional transmissions to and from the associated enodeB,repeater or CPE receiver. Transmissions do not only occur in a singledirection but involve both transmissions and receptions.

Referring now to FIG. 120, a base station 12002 utilizing a massive MIMOantenna configuration 11204 as described hereinabove transmits signalsbetween a relay/repeater 12006 using a link 12008 utilizing full duplexcommunications. Full-duplex communications utilize the same frequencyband and same time slots in the bidirectional communications between therelay/repeater 12006 and the base station 12002. For communicationsbetween the relay/repeater 12006 and a user terminal 12010, thecommunications link 12012 utilizes half duplex communications onspecific timeslots.

FIG. 121 illustrates the implementation of a millimeter waverelay/repeater within a transceiver dongle. The transceiver dongle 12102will connect to a USB port of a processing device using a USB interface12101. Signals are received and transmitted using an RF transceiver12106 as described herein. The signals can be converted between RF andbaseband using Baseband Processing Circuitry 12104 as described herein.Signal interference and bandwidth issues may be further overcome usingOAM processing circuitry 12108 that implements the various OAMprocessing techniques that are described herein above. The signals aretransmitted from and received at the dongle 12102 using an antenna array12110 that implemented any of the various configurations describedherein. OAM 12204 may be applied to the millimeter wave signals toimprove signal bandwidth and interference issues. MIMO 12206transmission techniques may be used to improve the transmission andreception characteristics of the millimeter wave signals. Artificialintelligence based digital cancellation techniques 12200 are used forlimiting interference between transmitted and received signals withinthe repeater/relay 12202 and self-interference caused by small distancesbetween antenna patches in mmWave system. Finally, full duplex channelestimation 12210 is used for controlling self-interference caused bysmall distances between antenna patches in mmWave system.

Referring now to FIG. 122, there are illustrated the variousfunctionalities and techniques implemented within a repeater/relay 12202implemented according to the present disclosure.

Thus, using the repeater/relay configurations discussed above utilizingthe describe manner for wireless signal communications described, andimprove manner for transmitting the signals and more particularlymillimeter wave signals may be provided.

It will be appreciated by those skilled in the art having the benefit ofthis disclosure that this hyper digital-analog mmwave repeater/relaywith full duplex system provides a unimproved matter for forwardingmillimeter wave signals from one point to another within a wirelesscommunication system. It should be understood that the drawings anddetailed description herein are to be regarded in an illustrative ratherthan a restrictive manner, and are not intended to be limiting to theparticular forms and examples disclosed. On the contrary, included areany further modifications, changes, rearrangements, substitutions,alternatives, design choices, and embodiments apparent to those ofordinary skill in the art, without departing from the spirit and scopehereof, as defined by the following claims. Thus, it is intended thatthe following claims be interpreted to embrace all such furthermodifications, changes, rearrangements, substitutions, alternatives,design choices, and embodiments.

What is claimed is:
 1. A system for transmitting millimeter wavesignals, comprising: a plurality of transceivers for communicating witha plurality of remote locations over millimeter wave communicationslinks, wherein each of the plurality of transceivers further comprises;a patch antenna array including a plurality of patch antennas, theplurality of patch antennas including a transmitter array portion in afirst orientation for transmitting signals and a receiver array portionin a second orientation for receiving signals, wherein the first andsecond orientations limit interference between the transmitted signalsand the received signals; baseband processing circuitry for convertingbetween millimeter wave signals and baseband signals; and wherein theplurality of transceivers relay the millimeter wave signals between atleast a first millimeter wave transceiver at first one of the pluralityof remote locations and a second millimeter wave transceiver at a secondone of the plurality of remote locations.
 2. The system of claim 1further comprising orbital angular momentum processing (OAM) processingcircuitry for applying a selected orbital angular momentum to thetransmitted signals being transmitted by a transceiver, wherein theselected orbital angular momentum prevents interference between thetransmitted signals and the received signals.
 3. The system of claim 1further comprising processing circuitry for providing artificialintelligence based digital cancellation of signal interference betweenthe transmitted signals and the received signals.
 4. The system of claim1 further comprising full duplex processing circuitry for providing fullduplex communications between the plurality of transceivers and theplurality of remote locations.
 5. The system of claim 1, wherein thetransmitter array portion comprises a first circular array of patchantennas and the receiver array portion comprises a second circulararray of patch antennas surrounding the first circular array of patchantennas, further wherein the second circular array of patch antennasand the first circular array of patch antennas are rotated at a selectedangle with respect to each other to limit interference between thetransmitted signals and the received signals.
 6. The system of claim 1,wherein the transmitter array portion comprises a first circular arrayof patch antennas and the receiver array portion comprises a secondcircular array of patch antennas surrounding the first circular array ofpatch antennas, further wherein an orientation of the plurality of patchantennas of the second circular array of patch antennas with respect tothe orientation of the plurality of patch antennas of the first circulararray of patch antennas is selected to limit interference between thetransmitted signals and the received signals.
 7. The system of claim 1,wherein the plurality of transceivers are implemented on a USB donglethat plugs into a USB port of a processing device.
 8. The system ofclaim 1, wherein the plurality of transceivers relay the millimeter wavesignals between at least a first millimeter wave transceiver at firstone of the plurality of remote locations and a plurality of millimeterwave transceivers at a plurality of the plurality of remote locations.9. A method for transmitting millimeter wave signals, comprising:communicating with a plurality of remote locations over millimeter wavecommunications links through a plurality of transceivers, wherein thestep of communicating further comprises; transmitting signals from atransmitter array portion of a patch antenna array including a pluralityof patch antennas; receiving signals on a receiver array portion of thepatch antenna array including the plurality of patch antennas; locatingthe transmitter array portion in a first orientation and the receiverarray portion in a second orientation to limit interference between thetransmitted signals and the received signals; converting betweenmillimeter wave signals and baseband signals using baseband processingcircuitry; and relaying the millimeter wave signals between at least afirst millimeter wave transceiver at first one of the plurality ofremote locations and a second millimeter wave transceiver at a secondone of the plurality of remote locations.
 10. The method of claim 9further comprising applying a selected orbital angular momentum to thetransmitted signals being transmitted by a transceiver using orbitalangular momentum (OAM) processing circuitry, wherein the selectedorbital angular momentum prevents interference between the transmittedsignals and the received signals.
 11. The method of claim 9 furthercomprising providing artificial intelligence based digital cancellationof signal interference between the transmitted signals and the receivedsignals.
 12. The method of claim 9 further comprising providing fullduplex communications between the plurality of transceivers and theplurality of remote locations using full duplex processing circuitry.13. The method of claim 9, wherein the step of locating furthercomprises rotating a second circular array of the patch antennas of thereceiver array portion and a first circular array of the patch antennasof the transmitter array portion at a selected angle with respect toeach other to limit interference between the transmitted signals and thereceived signals.
 14. The method of claim 13, wherein the step oflocating further comprises selecting the second orientation of theplurality of patch antennas of the second circular array with respect tothe first orientation of the plurality of patch antennas of the firstcircular array to limit interference between the transmitted signals andthe received signals.
 15. The method of claim 9 further comprising thestep of relaying the millimeter wave signals between at least a firstmillimeter wave transceiver at first one of the plurality of remotelocations and a plurality of millimeter wave transceivers at a pluralityof the plurality of remote locations.
 16. A system for transmittingmillimeter wave signals, comprising: a first millimeter wave transceiverfor transmitting and receiving millimeter wave signals from a firstremote location; a second millimeter wave transceiver for transmittingand receiving the millimeter wave signals from a second remote location;a plurality of transceivers for communicating with the first and thesecond millimeter wave transceivers over millimeter wave communicationslinks, wherein each of the plurality of transceivers further comprises;a patch antenna array including a plurality of patch antennas, theplurality of patch antennas including a transmitter array portion in afirst orientation for transmitting signals to the first and the secondmillimeter wave transceivers and a receiver array portion in a secondorientation for receiving signals from the first and the secondmillimeter wave transceivers, wherein the first and second orientationslimit interference between the transmitted signals and the receivedsignals; and wherein the plurality of transceivers relay the millimeterwave signals between at least the first millimeter wave transceiver atthe first remote location and the second millimeter wave transceiver atthe second remote location.
 17. The system of claim 16 furthercomprising orbital angular momentum processing (OAM) processingcircuitry for applying a selected orbital angular momentum to thetransmitted signals being transmitted by a transceiver, wherein theselected orbital angular momentum prevents interference between thetransmitted signals and the received signals.
 18. The system of claim 16further comprising processing circuitry for providing artificialintelligence based digital cancellation of signal interference betweenthe transmitted signals and the received signals.
 19. The system ofclaim 16 further comprising full duplex processing circuitry forproviding full duplex communications between the plurality oftransceivers.
 20. The system of claim 16, wherein the plurality oftransceivers are implemented on a USB dongle that plugs into a USB portof a processing device.
 21. The system of claim 16, wherein theplurality of transceivers comprise at least one of a Y-junctionrepeater, an H-junction repeater and a tandem Y-junction repeater. 22.The system of claim 16, wherein the plurality of transceivers comprise atandem repeater, the system further comprising a passive reflector forreflecting the transmitted signals from one of the plurality oftransceivers to the second millimeter wave transceiver.